Parallel processing of multiple channels with very narrow bandpass digital filtering

ABSTRACT

A method includes converting, by n analog to digital converter circuits, n analog signals into n first digital signals having a first data rate frequency; converting, by n digital decimation filtering circuits, the n first digital signals into n second digital signals having a second data rate frequency; and converting, by n digital bandpass filter (BPF) circuits, the n second digital signals into a plurality of outbound digital signals having a third data rate frequency. The coefficients for the taps of a digital BPF circuit is set to produce a bandpass region approximately centered at the oscillation frequency of the analog signal and having a bandwidth tuned for filtering a pure tone component of the analog signal. The first data rate frequency is a first integer multiple of the third data rate frequency. The second data rate frequency is a second integer multiple of the third data rate frequency.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present U.S. Utility Patent Application claims priority pursuant to35 U.S.C. § 120 as a continuation of U.S. Utility application Ser. No.16/780,133, entitled “ANALOG TO DIGITAL CONVERSION CIRCUIT WITH VERYNARROW BANDPASS DIGITAL FILTERING,” filed Feb. 3, 2020, issuing as U.S.Pat. No. 10,917,101 on Feb. 9, 2021, which is a continuation of U.S.Utility patent application Ser. No. 16/365,169 entitled “ANALOG TODIGITAL CONVERSION CIRCUIT WITH VERY NARROW BANDPASS DIGITAL FILTERING,”filed Mar. 26, 2019, issued as U.S. Pat. No. 10,554,215 on Feb. 4, 2020,both of which are hereby incorporated herein by reference in theirentirety and made part of the present U.S. Utility Patent Applicationfor all purposes.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

Not Applicable.

INCORPORATION-BY-REFERENCE OF MATERIAL SUBMITTED ON A COMPACT DISC

Not Applicable.

BACKGROUND OF THE INVENTION Technical Field of the Invention

This invention relates generally to data communication systems and moreparticularly to sensed data collection and/or communication.

Description of Related Art

Sensors are used in a wide variety of applications ranging from in-homeautomation, to industrial systems, to health care, to transportation,and so on. For example, sensors are placed in bodies, automobiles,airplanes, boats, ships, trucks, motorcycles, cell phones, televisions,touch-screens, industrial plants, appliances, motors, checkout counters,etc. for the variety of applications.

In general, a sensor converts a physical quantity into an electrical oroptical signal. For example, a sensor converts a physical phenomenon,such as a biological condition, a chemical condition, an electriccondition, an electromagnetic condition, a temperature, a magneticcondition, mechanical motion (position, velocity, acceleration, force,pressure), an optical condition, and/or a radioactivity condition, intoan electrical signal.

A sensor includes a transducer, which functions to convert one form ofenergy (e.g., force) into another form of energy (e.g., electricalsignal). There are a variety of transducers to support the variousapplications of sensors. For example, a transducer is capacitor, apiezoelectric transducer, a piezoresistive transducer, a thermaltransducer, a thermal-couple, a photoconductive transducer such as aphotoresistor, a photodiode, and/or phototransistor.

A sensor circuit is coupled to a sensor to provide the sensor with powerand to receive the signal representing the physical phenomenon from thesensor. The sensor circuit includes at least three electricalconnections to the sensor: one for a power supply; another for a commonvoltage reference (e.g., ground); and a third for receiving the signalrepresenting the physical phenomenon. The signal representing thephysical phenomenon will vary from the power supply voltage to ground asthe physical phenomenon changes from one extreme to another (for therange of sensing the physical phenomenon).

The sensor circuits provide the received sensor signals to one or morecomputing devices for processing. A computing device is known tocommunicate data, process data, and/or store data. The computing devicemay be a cellular phone, a laptop, a tablet, a personal computer (PC), awork station, a video game device, a server, and/or a data center thatsupport millions of web searches, stock trades, or on-line purchasesevery hour.

The computing device processes the sensor signals for a variety ofapplications. For example, the computing device processes sensor signalsto determine temperatures of a variety of items in a refrigerated truckduring transit. As another example, the computing device processes thesensor signals to determine a touch on a touch screen. As yet anotherexample, the computing device processes the sensor signals to determinevarious data points in a production line of a product.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING(S)

FIG. 1 is a schematic block diagram of an embodiment of a communicationsystem in accordance with the present invention;

FIG. 2 is a schematic block diagram of an embodiment of a computingdevice in accordance with the present invention;

FIG. 3 is a schematic block diagram of another embodiment of a computingdevice in accordance with the present invention;

FIG. 4 is a schematic block diagram of another embodiment of a computingdevice in accordance with the present invention;

FIG. 5A is a schematic plot diagram of a computing subsystem inaccordance with the present invention;

FIG. 5B is a schematic block diagram of another embodiment of acomputing subsystem in accordance with the present invention;

FIG. 5C is a schematic block diagram of another embodiment of acomputing subsystem in accordance with the present invention;

FIG. 5D is a schematic block diagram of another embodiment of acomputing subsystem in accordance with the present invention;

FIG. 5E is a schematic block diagram of another embodiment of acomputing subsystem in accordance with the present invention;

FIG. 6 is a schematic block diagram of an embodiment of a drive sensecircuit in accordance with the present invention;

FIG. 6A is a schematic block diagram of another embodiment of a drivesense circuit in accordance with the present invention;

FIG. 7 is an example of a power signal graph in accordance with thepresent invention;

FIG. 8 is an example of a sensor graph in accordance with the presentinvention;

FIG. 9 is a schematic block diagram of another example of a power signalgraph in accordance with the present invention;

FIG. 10 is a schematic block diagram of another example of a powersignal graph in accordance with the present invention;

FIG. 11 is a schematic block diagram of another example of a powersignal graph in accordance with the present invention;

FIG. 11A is a schematic block diagram of another example of a powersignal graph in accordance with the present invention;

FIG. 12 is a schematic block diagram of an embodiment of a power signalchange detection circuit in accordance with the present invention;

FIG. 13 is a schematic block diagram of another embodiment of a drivesense circuit in accordance with the present invention;

FIG. 14 is a schematic block diagram of an example of a drive sensecircuit with a programmed reference signal generator in accordance withthe present invention;

FIG. 15 is a schematic block diagram of an embodiment of a data sensingcircuit in accordance with the present invention;

FIG. 16 is a schematic block diagram of another embodiment of a datasensing circuit in accordance with the present invention;

FIG. 17 is a schematic block diagram of an embodiment of a data circuitin accordance with the present invention;

FIG. 18 is a schematic block diagram of an embodiment of an analog todigital conversion circuit in accordance with the present invention;

FIG. 19 is a schematic block diagram of another embodiment of an analogto digital conversion circuit in accordance with the present invention;

FIGS. 20A-20B are example graphs that plot condition verses capacitancein accordance with the present invention;

FIG. 21 is an example graph that plots impedance verses frequency for aninput in accordance with the present invention;

FIG. 22 is an example of affect values in accordance with the presentinvention;

FIG. 23 is a schematic block diagram of an embodiment of a sigma deltaanalog to digital (ADC) circuit in accordance with the presentinvention;

FIG. 24 an example of quantization noise of a sigma delta oversamplingmodulator in accordance with the present invention;

FIG. 25 is a schematic block diagram of example outputs of the differentstages of an analog to digital conversion circuit in accordance with thepresent invention;

FIG. 26 is an example of sampling an analog signal to produce adigitized signal in accordance with the present invention;

FIG. 27 is a schematic block diagram of a digital filter implementing amultiply-accumulate function in accordance with the present invention;

FIG. 28 is a schematic block diagram of a digital filter implementing amultiply-accumulate function in accordance with the present invention;

FIG. 29 is an example of a digitized signal in accordance with thepresent invention;

FIG. 30 is an example of producing a digital filtered output inaccordance with the present invention;

FIG. 31 is a schematic block diagram of an embodiment of a digitaldecimation filtering circuit in accordance with the present invention;

FIG. 32 is an example frequency response H(z) of an anti-aliasing filterin accordance with the present invention;

FIG. 33 is a schematic block diagram of an embodiment of ananti-aliasing filter in accordance with the present invention;

FIG. 34 is a schematic block diagram of an embodiment of a decimator inaccordance with the present invention;

FIG. 35 is an example of a frequency band having frequency channels inaccordance with the present invention;

FIG. 36 is a schematic block diagram of another embodiment of digitaldecimation filtering circuit in accordance with the present invention;

FIG. 37 is a schematic block diagram of another embodiment of digitaldecimation filtering circuit in accordance with the present invention;

FIG. 38 is a schematic block diagram of an example of polyphase filtersof digital decimation filtering circuit in accordance with the presentinvention;

FIG. 39 is a schematic block diagram of another embodiment of thedigital decimation filtering circuit in accordance with the presentinvention;

FIG. 40 is a schematic block diagram of an example of a shift registermemory in accordance with the present invention;

FIG. 41 is a schematic block diagram of another embodiment of thedigital decimation filtering circuit in accordance with the presentinvention;

FIG. 42 is an example of a frequency band having n frequency channels inaccordance with the present invention;

FIG. 43 is a schematic block diagram of an embodiment of digitalbandpass filter (BPF) circuit in accordance with the present invention;

FIG. 44 is an example frequency response H(z) of a digital bandpassfilter (BPF) circuit in accordance with the present invention;

FIG. 45 is an example frequency response H(z) of a digital bandpassfilter (BPF) circuit in accordance with the present invention;

FIGS. 46A-46D are examples of processing a signal by a digital bandpassfilter (BPF) circuit in accordance with the present invention;

FIGS. 47A-47D are examples of processing a signal by a digital bandpassfilter (BPF) circuit in accordance with the present invention;

FIG. 48 is a schematic block diagram of an embodiment of digitalbandpass filter (BPF) circuit in accordance with the present invention;

FIG. 49 is a schematic block diagram of another embodiment of analog todigital conversion circuit in accordance with the present invention;

FIG. 50 is a schematic block diagram of an embodiment of a firstbandpass filter (BPF) circuit in accordance with the present invention;

FIG. 51 is a schematic block diagram of an embodiment of a secondbandpass filter (BPF) circuit in accordance with the present invention;

FIG. 52 is a schematic block diagram of an embodiment of a coefficientprocessor in accordance with the present invention;

FIG. 53 is a schematic block diagram of another embodiment of analog todigital conversion circuit in accordance with the present invention; and

FIG. 54 is a schematic block diagram of an embodiment of processingmodule controls within the analog to digital conversion circuit.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 is a schematic block diagram of an embodiment of a communicationsystem 10 that includes a plurality of computing. devices 12-10, one ormore servers 22, one or more databases 24, one or more networks 26, aplurality of drive-sense circuits 28, a plurality of sensors 30, and aplurality of actuators 32. Computing devices 14 include a touch screen16 with sensors and drive-sensor circuits and computing devices 18include a touch & tactic screen 20 that includes sensors, actuators, anddrive-sense circuits.

A sensor 30 functions to convert a physical input into an electricaloutput and/or an optical output. The physical input of a sensor may beone of a variety of physical input conditions. For example, the physicalcondition includes one or more of, but is not limited to, acoustic waves(e.g., amplitude, phase, polarization, spectrum, and/or wave velocity);a biological and/or chemical condition (e.g., fluid concentration,level, composition, etc.); an electric condition (e.g., charge, voltage,current, conductivity, permittivity, eclectic field, which includesamplitude, phase, and/or polarization); a magnetic condition (e.g.,flux, permeability, magnetic field, which amplitude, phase, and/orpolarization); an optical condition (e.g., refractive index,reflectivity, absorption, etc.); a thermal condition (e.g., temperature,flux, specific heat, thermal conductivity, etc.); and a mechanicalcondition (e.g., position, velocity, acceleration, force, strain,stress, pressure, torque, etc.). For example, piezoelectric sensorconverts force or pressure into an eclectic signal. As another example,a microphone converts audible acoustic waves into electrical signals.

There are a variety of types of sensors to sense the various types ofphysical conditions. Sensor types include, but are not limited to,capacitor sensors, inductive sensors, accelerometers, piezoelectricsensors, light sensors, magnetic field sensors, ultrasonic sensors,temperature sensors, infrared (IR) sensors, touch sensors, proximitysensors, pressure sensors, level sensors, smoke sensors, and gassensors. In many ways, sensors function as the interface between thephysical world and the digital world by converting real world conditionsinto digital signals that are then processed by computing devices for avast number of applications including, but not limited to, medicalapplications, production automation applications, home environmentcontrol, public safety, and so on.

The various types of sensors have a variety of sensor characteristicsthat are factors in providing power to the sensors, receiving signalsfrom the sensors, and/or interpreting the signals from the sensors. Thesensor characteristics include resistance, reactance, powerrequirements, sensitivity, range, stability, repeatability, linearity,error, response time, and/or frequency response. For example, theresistance, reactance, and/or power requirements are factors indetermining drive circuit requirements. As another example, sensitivity,stability, and/or linear are factors for interpreting the measure of thephysical condition based on the received electrical and/or opticalsignal (e.g., measure of temperature, pressure, etc.).

An actuator 32 converts an electrical input into a physical output. Thephysical output of an actuator may be one of a variety of physicaloutput conditions. For example, the physical output condition includesone or more of, but is not limited to, acoustic waves (e.g., amplitude,phase, polarization, spectrum, and/or wave velocity); a magneticcondition (e.g., flux, permeability, magnetic field, which amplitude,phase, and/or polarization); a thermal condition (e.g., temperature,flux, specific heat, thermal conductivity, etc.); and a mechanicalcondition (e.g., position, velocity, acceleration, force, strain,stress, pressure, torque, etc.). As an example, a piezoelectric actuatorconverts voltage into force or pressure. As another example, a speakerconverts electrical signals into audible acoustic waves.

An actuator 32 may be one of a variety of actuators. For example, anactuator 32 is one of a comb drive, a digital micro-mirror device, anelectric motor, an electroactive polymer, a hydraulic cylinder, apiezoelectric actuator, a pneumatic actuator, a screw jack, aservomechanism, a solenoid, a stepper motor, a shape-memory allow, athermal bimorph, and a hydraulic actuator.

The various types of actuators have a variety of actuatorscharacteristics that are factors in providing power to the actuator andsending signals to the actuators for desired performance. The actuatorcharacteristics include resistance, reactance, power requirements,sensitivity, range, stability, repeatability, linearity, error, responsetime, and/or frequency response. For example, the resistance, reactance,and power requirements are factors in determining drive circuitrequirements. As another example, sensitivity, stability, and/or linearare factors for generating the signaling to send to the actuator toobtain the desired physical output condition.

The computing devices 12, 14, and 18 may each be a portable computingdevice and/or a fixed computing device. A portable computing device maybe a social networking device, a gaming device, a cell phone, a smartphone, a digital assistant, a digital music player, a digital videoplayer, a laptop computer, a handheld computer, a tablet, a video gamecontroller, and/or any other portable device that includes a computingcore. A fixed computing device may be a computer (PC), a computerserver, a cable set-top box, a satellite receiver, a television set, aprinter, a fax machine, home entertainment equipment, a video gameconsole, and/or any type of home or office computing equipment. Thecomputing devices 12, 14, and 18 will be discussed in greater detailwith reference to one or more of FIGS. 2-4.

A server 22 is a special type of computing device that is optimized forprocessing large amounts of data requests in parallel. A server 22includes similar components to that of the computing devices 12, 14,and/or 18 with more robust processing modules, more main memory, and/ormore hard drive memory (e.g., solid state, hard drives, etc.). Further,a server 22 is typically accessed remotely; as such it does notgenerally include user input devices and/or user output devices. Inaddition, a server may be a standalone separate computing device and/ormay be a cloud computing device.

A database 24 is a special type of computing device that is optimizedfor large scale data storage and retrieval. A database 24 includessimilar components to that of the computing devices 12, 14, and/or 18with more hard drive memory (e.g., solid state, hard drives, etc.) andpotentially with more processing modules and/or main memory. Further, adatabase 24 is typically accessed remotely; as such it does notgenerally include user input devices and/or user output devices. Inaddition, a database 24 may be a standalone separate computing deviceand/or may be a cloud computing device.

The network 26 includes one more local area networks (LAN) and/or one ormore wide area networks WAN), which may be a public network and/or aprivate network. A LAN may be a wireless-LAN (e.g., Wi-Fi access point,Bluetooth, ZigBee, etc.) and/or a wired network (e.g., Firewire,Ethernet, etc.). A WAN may be a wired and/or wireless WAN. For example,a LAN may be a personal home or business's wireless network and a WAN isthe Internet, cellular telephone infrastructure, and/or satellitecommunication infrastructure.

In an example of operation, computing device 12-1 communicates with aplurality of drive-sense circuits 28, which, in turn, communicate with aplurality of sensors 30. The sensors 30 and/or the drive-sense circuits28 are within the computing device 12-1 and/or external to it. Forexample, the sensors 30 may be external to the computing device 12-1 andthe drive-sense circuits are within the computing device 12-1. Asanother example, both the sensors 30 and the drive-sense circuits 28 areexternal to the computing device 12-1. When the drive-sense circuits 28are external to the computing device, they are coupled to the computingdevice 12-1 via wired and/or wireless communication links as will bediscussed in greater detail with reference to one or more of FIGS.5A-5C.

The computing device 12-1 communicates with the drive-sense circuits 28to; (a) turn them on, (b) obtain data from the sensors (individuallyand/or collectively), (c) instruct the drive sense circuit on how tocommunicate the sensed data to the computing device 12-1, (d) providesignaling attributes (e.g., DC level, AC level, frequency, power level,regulated current signal, regulated voltage signal, regulation of animpedance, frequency patterns for various sensors, different frequenciesfor different sensing applications, etc.) to use with the sensors,and/or (e) provide other commands and/or instructions.

As a specific example, the sensors 30 are distributed along a pipelineto measure flow rate and/or pressure within a section of the pipeline.The drive-sense circuits 28 have their own power source (e.g., battery,power supply, etc.) and are proximally located to their respectivesensors 30. At desired time intervals (milliseconds, seconds, minutes,hours, etc.), the drive-sense circuits 28 provide a regulated sourcesignal or a power signal to the sensors 30. An electrical characteristicof the sensor 30 affects the regulated source signal or power signal,which is reflective of the condition (e.g., the flow rate and/or thepressure) that sensor is sensing.

The drive-sense circuits 28 detect the effects on the regulated sourcesignal or power signals as a result of the electrical characteristics ofthe sensors. The drive-sense circuits 28 then generate signalsrepresentative of change to the regulated source signal or power signalbased on the detected effects on the power signals. The changes to theregulated source signals or power signals are representative of theconditions being sensed by the sensors 30.

The drive-sense circuits 28 provide the representative signals of theconditions to the computing device 12-1. A representative signal may bean analog signal or a digital signal. In either case, the computingdevice 12-1 interprets the representative signals to determine thepressure and/or flow rate at each sensor location along the pipeline.The computing device may then provide this information to the server 22,the database 24, and/or to another computing device for storing and/orfurther processing.

As another example of operation, computing device 12-2 is coupled to adrive-sense circuit 28, which is, in turn, coupled to a senor 30. Thesensor 30 and/or the drive-sense circuit 28 may be internal and/orexternal to the computing device 12-2. In this example, the sensor 30 issensing a condition that is particular to the computing device 12-2. Forexample, the sensor 30 may be a temperature sensor, an ambient lightsensor, an ambient noise sensor, etc. As described above, wheninstructed by the computing device 12-2 (which may be a default settingfor continuous sensing or at regular intervals), the drive-sense circuit28 provides the regulated source signal or power signal to the sensor 30and detects an effect to the regulated source signal or power signalbased on an electrical characteristic of the sensor. The drive-sensecircuit generates a representative signal of the affect and sends it tothe computing device 12-2.

In another example of operation, computing device 12-3 is coupled to aplurality of drive-sense circuits 28 that are coupled to a plurality ofsensors 30 and is coupled to a plurality of drive-sense circuits 28 thatare coupled to a plurality of actuators 32. The generally functionalityof the drive-sense circuits 28 coupled to the sensors 30 in accordancewith the above description.

Since an actuator 32 is essentially an inverse of a sensor in that anactuator converts an electrical signal into a physical condition, whilea sensor converts a physical condition into an electrical signal, thedrive-sense circuits 28 can be used to power actuators 32. Thus, in thisexample, the computing device 12-3 provides actuation signals to thedrive-sense circuits 28 for the actuators 32. The drive-sense circuitsmodulate the actuation signals on to power signals or regulated controlsignals, which are provided to the actuators 32. The actuators 32 arepowered from the power signals or regulated control signals and producethe desired physical condition from the modulated actuation signals.

As another example of operation, computing device 12-x is coupled to adrive-sense circuit 28 that is coupled to a sensor 30 and is coupled toa drive-sense circuit 28 that is coupled to an actuator 32. In thisexample, the sensor 30 and the actuator 32 are for use by the computingdevice 12-x. For example, the sensor 30 may be a piezoelectricmicrophone and the actuator 32 may be a piezoelectric speaker.

FIG. 2 is a schematic block diagram of an embodiment of a computingdevice 12 (e.g., any one of 12-1 through 12-x). The computing device 12includes a core control module 40, one or more processing modules 42,one or more main memories 44, cache memory 46, a video graphicsprocessing module 48, a display 50, an Input-Output (I/O) peripheralcontrol module 52, one or more input interface modules 56, one or moreoutput interface modules 58, one or more network interface modules 60,and one or more memory interface modules 62. A processing module 42 isdescribed in greater detail at the end of the detailed description ofthe invention section and, in an alternative embodiment, has a directionconnection to the main memory 44. In an alternate embodiment, the corecontrol module 40 and the I/O and/or peripheral control module 52 areone module, such as a chipset, a quick path interconnect (QPI), and/oran ultra-path interconnect (UPI).

Each of the main memories 44 includes one or more Random Access Memory(RAM) integrated circuits, or chips. For example, a main memory 44includes four DDR4 (4th generation of double data rate) RAM chips, eachrunning at a rate of 2,400 MHz. In general, the main memory 44 storesdata and operational instructions most relevant for the processingmodule 42. For example, the core control module 40 coordinates thetransfer of data and/or operational instructions from the main memory 44and the memory 64-66. The data and/or operational instructions retrievefrom memory 64-66 are the data and/or operational instructions requestedby the processing module or will most likely be needed by the processingmodule. When the processing module is done with the data and/oroperational instructions in main memory, the core control module 40coordinates sending updated data to the memory 64-66 for storage.

The memory 64-66 includes one or more hard drives, one or more solidstate memory chips, and/or one or more other large capacity storagedevices that, in comparison to cache memory and main memory devices,is/are relatively inexpensive with respect to cost per amount of datastored. The memory 64-66 is coupled to the core control module 40 viathe I/O and/or peripheral control module 52 and via one or more memoryinterface modules 62. In an embodiment, the I/O and/or peripheralcontrol module 52 includes one or more Peripheral Component Interface(PCI) buses to which peripheral components connect to the core controlmodule 40. A memory interface module 62 includes a software driver and ahardware connector for coupling a memory device to the I/O and/orperipheral control module 52. For example, a memory interface 62 is inaccordance with a Serial Advanced Technology Attachment (SATA) port.

The core control module 40 coordinates data communications between theprocessing module(s) 42 and the network(s) 26 via the I/O and/orperipheral control module 52, the network interface module(s) 60, and anetwork card 68 or 70. A network card 68 or 70 includes a wirelesscommunication unit or a wired communication unit. A wirelesscommunication unit includes a wireless local area network (WLAN)communication device, a cellular communication device, a Bluetoothdevice, and/or a ZigBee communication device. A wired communication unitincludes a Gigabit LAN connection, a Firewire connection, and/or aproprietary computer wired connection. A network interface module 60includes a software driver and a hardware connector for coupling thenetwork card to the I/O and/or peripheral control module 52. Forexample, the network interface module 60 is in accordance with one ormore versions of IEEE 802.11, cellular telephone protocols, 10/100/1000Gigabit LAN protocols, etc.

The core control module 40 coordinates data communications between theprocessing module(s) 42 and input device(s) 72 via the input interfacemodule(s) 56 and the I/O and/or peripheral control module 52. An inputdevice 72 includes a keypad, a keyboard, control switches, a touchpad, amicrophone, a camera, etc. An input interface module 56 includes asoftware driver and a hardware connector for coupling an input device tothe I/O and/or peripheral control module 52. In an embodiment, an inputinterface module 56 is in accordance with one or more Universal SerialBus (USB) protocols.

The core control module 40 coordinates data communications between theprocessing module(s) 42 and output device(s) 74 via the output interfacemodule(s) 58 and the I/O and/or peripheral control module 52. An outputdevice 74 includes a speaker, etc. An output interface module 58includes a software driver and a hardware connector for coupling anoutput device to the I/O and/or peripheral control module 52. In anembodiment, an output interface module 56 is in accordance with one ormore audio codec protocols.

The processing module 42 communicates directly with a video graphicsprocessing module 48 to display data on the display 50. The display 50includes an LED (light emitting diode) display, an LCD (liquid crystaldisplay), and/or other type of display technology. The display has aresolution, an aspect ratio, and other features that affect the qualityof the display. The video graphics processing module 48 receives datafrom the processing module 42, processes the data to produce rendereddata in accordance with the characteristics of the display, and providesthe rendered data to the display 50.

FIG. 2 further illustrates sensors 30 and actuators 32 coupled todrive-sense circuits 28, which are coupled to the input interface module56 (e.g., USB port). Alternatively, one or more of the drive-sensecircuits 28 is coupled to the computing device via a wireless networkcard (e.g., WLAN) or a wired network card (e.g., Gigabit LAN). While notshown, the computing device 12 further includes a BIOS (Basic InputOutput System) memory coupled to the core control module 40.

FIG. 3 is a schematic block diagram of another embodiment of a computingdevice 14 that includes a core control module 40, one or more processingmodules 42, one or more main memories 44, cache memory 46, a videographics processing module 48, a touch screen 16, an Input-Output (I/O)peripheral control module 52, one or more input interface modules 56,one or more output interface modules 58, one or more network interfacemodules 60, and one or more memory interface modules 62. The touchscreen 16 includes a touch screen display 80, a plurality of sensors 30,a plurality of drive-sense circuits (DSC), and a touch screen processingmodule 82.

Computing device 14 operates similarly to computing device 12 of FIG. 2with the addition of a touch screen as an input device. The touch screenincludes a plurality of sensors (e.g., electrodes, capacitor sensingcells, capacitor sensors, inductive sensor, etc.) to detect a proximaltouch of the screen. For example, when one or more fingers touches thescreen, capacitance of sensors proximal to the touch(es) are affected(e.g., impedance changes). The drive-sense circuits (DSC) coupled to theaffected sensors detect the change and provide a representation of thechange to the touch screen processing module 82, which may be a separateprocessing module or integrated into the processing module 42.

The touch screen processing module 82 processes the representativesignals from the drive-sense circuits (DSC) to determine the location ofthe touch(es). This information is inputted to the processing module 42for processing as an input. For example, a touch represents a selectionof a button on screen, a scroll function, a zoom in-out function, etc.

FIG. 4 is a schematic block diagram of another embodiment of a computingdevice 18 that includes a core control module 40, one or more processingmodules 42, one or more main memories 44, cache memory 46, a videographics processing module 48, a touch and tactile screen 20, anInput-Output (I/O) peripheral control module 52, one or more inputinterface modules 56, one or more output interface modules 58, one ormore network interface modules 60, and one or more memory interfacemodules 62. The touch and tactile screen 20 includes a touch and tactilescreen display 90, a plurality of sensors 30, a plurality of actuators32, a plurality of drive-sense circuits (DSC), a touch screen processingmodule 82, and a tactile screen processing module 92.

Computing device 18 operates similarly to computing device 14 of FIG. 3with the addition of a tactile aspect to the screen 20 as an outputdevice. The tactile portion of the screen 20 includes the plurality ofactuators (e.g., piezoelectric transducers to create vibrations,solenoids to create movement, etc.) to provide a tactile feel to thescreen 20. To do so, the processing module creates tactile data, whichis provided to the appropriate drive-sense circuits (DSC) via thetactile screen processing module 92, which may be a stand-aloneprocessing module or integrated into processing module 42. Thedrive-sense circuits (DSC) convert the tactile data into drive-actuatesignals and provide them to the appropriate actuators to create thedesired tactile feel on the screen 20.

FIG. 5A is a schematic plot diagram of a computing subsystem 25 thatincludes a sensed data processing module 65, a plurality ofcommunication modules 61A-x, a plurality of processing modules 42A-x, aplurality of drive sense circuits 28, and a plurality of sensors 1-x,which may be sensors 30 of FIG. 1. The sensed data processing module 65is one or more processing modules within one or more servers 22 and/orone more processing modules in one or more computing devices that aredifferent than the computing devices in which processing modules 42A-xreside.

A drive-sense circuit 28 (or multiple drive-sense circuits), aprocessing module (e.g., 41A), and a communication module (e.g., 61A)are within a common computing device. Each grouping of a drive-sensecircuit(s), processing module, and communication module is in a separatecomputing device. A communication module 61A-x is constructed inaccordance with one or more wired communication protocol and/or one ormore wireless communication protocols that is/are in accordance with theone or more of the Open System Interconnection (OSI) model, theTransmission Control Protocol/Internet Protocol (TCP/IP) model, andother communication protocol module.

In an example of operation, a processing module (e.g., 42A) provides acontrol signal to its corresponding drive-sense circuit 28. Theprocessing module 42 A may generate the control signal, receive it fromthe sensed data processing module 65, or receive an indication from thesensed data processing module 65 to generate the control signal. Thecontrol signal enables the drive-sense circuit 28 to provide a drivesignal to its corresponding sensor. The control signal may furtherinclude a reference signal having one or more frequency components tofacilitate creation of the drive signal and/or interpreting a sensedsignal received from the sensor.

Based on the control signal, the drive-sense circuit 28 provides thedrive signal to its corresponding sensor (e.g., 1) on a drive & senseline. While receiving the drive signal (e.g., a power signal, aregulated source signal, etc.), the sensor senses a physical condition1-x (e.g., acoustic waves, a biological condition, a chemical condition,an electric condition, a magnetic condition, an optical condition, athermal condition, and/or a mechanical condition). As a result of thephysical condition, an electrical characteristic (e.g., impedance,voltage, current, capacitance, inductance, resistance, reactance, etc.)of the sensor changes, which affects the drive signal. Note that if thesensor is an optical sensor, it converts a sensed optical condition intoan electrical characteristic.

The drive-sense circuit 28 detects the effect on the drive signal viathe drive & sense line and processes the affect to produce a signalrepresentative of power change, which may be an analog or digitalsignal. The processing module 42A receives the signal representative ofpower change, interprets it, and generates a value representing thesensed physical condition. For example, if the sensor is sensingpressure, the value representing the sensed physical condition is ameasure of pressure (e.g., x PSI (pounds per square inch)).

In accordance with a sensed data process function (e.g., algorithm,application, etc.), the sensed data processing module 65 gathers thevalues representing the sensed physical conditions from the processingmodules. Since the sensors 1-x may be the same type of sensor (e.g., apressure sensor), may each be different sensors, or a combinationthereof; the sensed physical conditions may be the same, may each bedifferent, or a combination thereof. The sensed data processing module65 processes the gathered values to produce one or more desired results.For example, if the computing subsystem 25 is monitoring pressure alonga pipeline, the processing of the gathered values indicates that thepressures are all within normal limits or that one or more of the sensedpressures is not within normal limits.

As another example, if the computing subsystem 25 is used in amanufacturing facility, the sensors are sensing a variety of physicalconditions, such as acoustic waves (e.g., for sound proofing, soundgeneration, ultrasound monitoring, etc.), a biological condition (e.g.,a bacterial contamination, etc.) a chemical condition (e.g.,composition, gas concentration, etc.), an electric condition (e.g.,current levels, voltage levels, electro-magnetic interference, etc.), amagnetic condition (e.g., induced current, magnetic field strength,magnetic field orientation, etc.), an optical condition (e.g., ambientlight, infrared, etc.), a thermal condition (e.g., temperature, etc.),and/or a mechanical condition (e.g., physical position, force, pressure,acceleration, etc.).

The computing subsystem 25 may further include one or more actuators inplace of one or more of the sensors and/or in addition to the sensors.When the computing subsystem 25 includes an actuator, the correspondingprocessing module provides an actuation control signal to thecorresponding drive-sense circuit 28. The actuation control signalenables the drive-sense circuit 28 to provide a drive signal to theactuator via a drive & actuate line (e.g., similar to the drive & senseline, but for the actuator). The drive signal includes one or morefrequency components and/or amplitude components to facilitate a desiredactuation of the actuator.

In addition, the computing subsystem 25 may include an actuator andsensor working in concert. For example, the sensor is sensing thephysical condition of the actuator. In this example, a drive-sensecircuit provides a drive signal to the actuator and another drive sensesignal provides the same drive signal, or a scaled version of it, to thesensor. This allows the sensor to provide near immediate and continuoussensing of the actuator's physical condition. This further allows forthe sensor to operate at a first frequency and the actuator to operateat a second frequency.

In an embodiment, the computing subsystem is a stand-alone system for awide variety of applications (e.g., manufacturing, pipelines, testing,monitoring, security, etc.). In another embodiment, the computingsubsystem 25 is one subsystem of a plurality of subsystems forming alarger system. For example, different subsystems are employed based ongeographic location. As a specific example, the computing subsystem 25is deployed in one section of a factory and another computing subsystemis deployed in another part of the factory. As another example,different subsystems are employed based function of the subsystems. As aspecific example, one subsystem monitors a city's traffic lightoperation and another subsystem monitors the city's sewage treatmentplants.

Regardless of the use and/or deployment of the computing system, thephysical conditions it is sensing, and/or the physical conditions it isactuating, each sensor and each actuator (if included) is driven andsensed by a single line as opposed to separate drive and sense lines.This provides many advantages including, but not limited to, lower powerrequirements, better ability to drive high impedance sensors, lower lineto line interference, and/or concurrent sensing functions.

FIG. 5B is a schematic block diagram of another embodiment of acomputing subsystem 25 that includes a sensed data processing module 65,a communication module 61, a plurality of processing modules 42A-x, aplurality of drive sense circuits 28, and a plurality of sensors 1-x,which may be sensors 30 of FIG. 1. The sensed data processing module 65is one or more processing modules within one or more servers 22 and/orone more processing modules in one or more computing devices that aredifferent than the computing device, devices, in which processingmodules 42A-x reside.

In an embodiment, the drive-sense circuits 28, the processing modules,and the communication module are within a common computing device. Forexample, the computing device includes a central processing unit thatincludes a plurality of processing modules. The functionality andoperation of the sensed data processing module 65, the communicationmodule 61, the processing modules 42A-x, the drive sense circuits 28,and the sensors 1-x are as discussed with reference to FIG. 5A.

FIG. 5C is a schematic block diagram of another embodiment of acomputing subsystem 25 that includes a sensed data processing module 65,a communication module 61, a processing module 42, a plurality of drivesense circuits 28, and a plurality of sensors 1-x, which may be sensors30 of FIG. 1. The sensed data processing module 65 is one or moreprocessing modules within one or more servers 22 and/or one moreprocessing modules in one or more computing devices that are differentthan the computing device in which the processing module 42 resides.

In an embodiment, the drive-sense circuits 28, the processing module,and the communication module are within a common computing device. Thefunctionality and operation of the sensed data processing module 65, thecommunication module 61, the processing module 42, the drive sensecircuits 28, and the sensors 1-x are as discussed with reference to FIG.5A.

FIG. 5D is a schematic block diagram of another embodiment of acomputing subsystem 25 that includes a processing module 42, a referencesignal circuit 100, a plurality of drive sense circuits 28, and aplurality of sensors 30. The processing module 42 includes a drive-senseprocessing block 104, a drive-sense control block 102, and a referencecontrol block 106. Each block 102-106 of the processing module 42 may beimplemented via separate modules of the processing module, may be acombination of software and hardware within the processing module,and/or may be field programmable modules within the processing module42.

In an example of operation, the drive-sense control block 104 generatesone or more control signals to activate one or more of the drive-sensecircuits 28. For example, the drive-sense control block 102 generates acontrol signal that enables of the drive-sense circuits 28 for a givenperiod of time (e.g., 1 second, 1 minute, etc.). As another example, thedrive-sense control block 102 generates control signals to sequentiallyenable the drive-sense circuits 28. As yet another example, thedrive-sense control block 102 generates a series of control signals toperiodically enable the drive-sense circuits 28 (e.g., enabled onceevery second, every minute, every hour, etc.).

Continuing with the example of operation, the reference control block106 generates a reference control signal that it provides to thereference signal circuit 100. The reference signal circuit 100generates, in accordance with the control signal, one or more referencesignals for the drive-sense circuits 28. For example, the control signalis an enable signal, which, in response, the reference signal circuit100 generates a pre-programmed reference signal that it provides to thedrive-sense circuits 28. In another example, the reference signalcircuit 100 generates a unique reference signal for each of thedrive-sense circuits 28. In yet another example, the reference signalcircuit 100 generates a first unique reference signal for each of thedrive-sense circuits 28 in a first group and generates a second uniquereference signal for each of the drive-sense circuits 28 in a secondgroup.

The reference signal circuit 100 may be implemented in a variety ofways. For example, the reference signal circuit 100 includes a DC(direct current) voltage generator, an AC voltage generator, and avoltage combining circuit. The DC voltage generator generates a DCvoltage at a first level and the AC voltage generator generates an ACvoltage at a second level, which is less than or equal to the firstlevel. The voltage combining circuit combines the DC and AC voltages toproduce the reference signal. As examples, the reference signal circuit100 generates a reference signal similar to the signals shown in FIG. 7,which will be subsequently discussed.

As another example, the reference signal circuit 100 includes a DCcurrent generator, an AC current generator, and a current combiningcircuit. The DC current generator generates a DC current a first currentlevel and the AC current generator generates an AC current at a secondcurrent level, which is less than or equal to the first current level.The current combining circuit combines the DC and AC currents to producethe reference signal.

Returning to the example of operation, the reference signal circuit 100provides the reference signal, or signals, to the drive-sense circuits28. When a drive-sense circuit 28 is enabled via a control signal fromthe drive sense control block 102, it provides a drive signal to itscorresponding sensor 30. As a result of a physical condition, anelectrical characteristic of the sensor is changed, which affects thedrive signal. Based on the detected effect on the drive signal and thereference signal, the drive-sense circuit 28 generates a signalrepresentative of the effect on the drive signal.

The drive-sense circuit provides the signal representative of the effecton the drive signal to the drive-sense processing block 104. Thedrive-sense processing block 104 processes the representative signal toproduce a sensed value 97 of the physical condition (e.g., a digitalvalue that represents a specific temperature, a specific pressure level,etc.). The processing module 42 provides the sensed value 97 to anotherapplication running on the computing device, to another computingdevice, and/or to a server 22.

FIG. 5E is a schematic block diagram of another embodiment of acomputing subsystem 25 that includes a processing module 42, a pluralityof drive sense circuits 28, and a plurality of sensors 30. Thisembodiment is similar to the embodiment of FIG. 5D with thefunctionality of the drive-sense processing block 104, a drive-sensecontrol block 102, and a reference control block 106 shown in greaterdetail. For instance, the drive-sense control block 102 includesindividual enable/disable blocks 102-1 through 102-y. An enable/disableblock functions to enable or disable a corresponding drive-sense circuitin a manner as discussed above with reference to FIG. 5D.

The drive-sense processing block 104 includes variance determiningmodules 104-1 a through y and variance interpreting modules 104-2 athrough y. For example, variance determining module 104-1 a receives,from the corresponding drive-sense circuit 28, a signal representativeof a physical condition sensed by a sensor. The variance determiningmodule 104-1 a functions to determine a difference from the signalrepresenting the sensed physical condition with a signal representing aknown, or reference, physical condition. The variance interpretingmodule 104-1 b interprets the difference to determine a specific valuefor the sensed physical condition.

As a specific example, the variance determining module 104-1 a receivesa digital signal of 1001 0110 (150 in decimal) that is representative ofa sensed physical condition (e.g., temperature) sensed by a sensor fromthe corresponding drive-sense circuit 28. With 8-bits, there are 28(256) possible signals representing the sensed physical condition.Assume that the units for temperature is Celsius and a digital value of0100 0000 (64 in decimal) represents the known value for 25 degreeCelsius. The variance determining module 104-b 1 determines thedifference between the digital signal representing the sensed value(e.g., 1001 0110, 150 in decimal) and the known signal value of (e.g.,0100 0000, 64 in decimal), which is 0011 0000 (86 in decimal). Thevariance determining module 104-b 1 then determines the sensed valuebased on the difference and the known value. In this example, the sensedvalue equals 25+86*(100/256)=25+33.6=58.6 degrees Celsius.

FIG. 6 is a schematic block diagram of a drive center circuit 28-acoupled to a sensor 30. The drive sense-sense circuit 28 includes apower source circuit 110 and a power signal change detection circuit112. The sensor 30 includes one or more transducers that have varyingelectrical characteristics (e.g., capacitance, inductance, impedance,current, voltage, etc.) based on varying physical conditions 114 (e.g.,pressure, temperature, biological, chemical, etc.), or vice versa (e.g.,an actuator).

The power source circuit 110 is operably coupled to the sensor 30 and,when enabled (e.g., from a control signal from the processing module 42,power is applied, a switch is closed, a reference signal is received,etc.) provides a power signal 116 to the sensor 30. The power sourcecircuit 110 may be a voltage supply circuit (e.g., a battery, a linearregulator, an unregulated DC-to-DC converter, etc.) to produce avoltage-based power signal, a current supply circuit (e.g., a currentsource circuit, a current mirror circuit, etc.) to produce acurrent-based power signal, or a circuit that provide a desired powerlevel to the sensor and substantially matches impedance of the sensor.The power source circuit 110 generates the power signal 116 to include aDC (direct current) component and/or an oscillating component.

When receiving the power signal 116 and when exposed to a condition 114,an electrical characteristic of the sensor affects 118 the power signal.When the power signal change detection circuit 112 is enabled, itdetects the affect 118 on the power signal as a result of the electricalcharacteristic of the sensor. For example, the power signal is a 1.5voltage signal, and, under a first condition, the sensor draws 1milliamp of current, which corresponds to an impedance of 1.5 K Ohms.Under a second conditions, the power signal remains at 1.5 volts and thecurrent increases to 1.5 milliamps. As such, from condition 1 tocondition 2, the impedance of the sensor changed from 1.5 K Ohms to 1 KOhms. The power signal change detection circuit 112 determines thischange and generates a representative signal 120 of the change to thepower signal.

As another example, the power signal is a 1.5 voltage signal, and, undera first condition, the sensor draws 1 milliamp of current, whichcorresponds to an impedance of 1.5 K Ohms. Under a second conditions,the power signal drops to 1.3 volts and the current increases to 1.3milliamps. As such, from condition 1 to condition 2, the impedance ofthe sensor changed from 1.5 K Ohms to 1 K Ohms. The power signal changedetection circuit 112 determines this change and generates arepresentative signal 120 of the change to the power signal.

The power signal 116 includes a DC component 122 and/or an oscillatingcomponent 124 as shown in FIG. 7. The oscillating component 124 includesa sinusoidal signal, a square wave signal, a triangular wave signal, amultiple level signal (e.g., has varying magnitude over time withrespect to the DC component), and/or a polygonal signal (e.g., has asymmetrical or asymmetrical polygonal shape with respect to the DCcomponent). Note that the power signal is shown without affect from thesensor as the result of a condition or changing condition.

In an embodiment, power generating circuit 110 varies frequency of theoscillating component 124 of the power signal 116 so that it can betuned to the impedance of the sensor and/or to be off-set in frequencyfrom other power signals in a system. For example, a capacitancesensor's impedance decreases with frequency. As such, if the frequencyof the oscillating component is too high with respect to thecapacitance, the capacitor looks like a short and variances incapacitances will be missed. Similarly, if the frequency of theoscillating component is too low with respect to the capacitance, thecapacitor looks like an open and variances in capacitances will bemissed.

In an embodiment, the power generating circuit 110 varies magnitude ofthe DC component 122 and/or the oscillating component 124 to improveresolution of sensing and/or to adjust power consumption of sensing. Inaddition, the power generating circuit 110 generates the drive signal110 such that the magnitude of the oscillating component 124 is lessthan magnitude of the DC component 122.

FIG. 6A is a schematic block diagram of a drive center circuit 28-a 1coupled to a sensor 30. The drive sense-sense circuit 28-a 1 includes asignal source circuit 111, a signal change detection circuit 113, and apower source 115. The power source 115 (e.g., a battery, a power supply,a current source, etc.) generates a voltage and/or current that iscombined with a signal 117, which is produced by the signal sourcecircuit 111. The combined signal is supplied to the sensor 30.

The signal source circuit 111 may be a voltage supply circuit (e.g., abattery, a linear regulator, an unregulated DC-to-DC converter, etc.) toproduce a voltage-based signal 117, a current supply circuit (e.g., acurrent source circuit, a current mirror circuit, etc.) to produce acurrent-based signal 117, or a circuit that provide a desired powerlevel to the sensor and substantially matches impedance of the sensor.The signal source circuit 111 generates the signal 117 to include a DC(direct current) component and/or an oscillating component.

When receiving the combined signal (e.g., signal 117 and power from thepower source) and when exposed to a condition 114, an electricalcharacteristic of the sensor affects 119 the signal. When the signalchange detection circuit 113 is enabled, it detects the affect 119 onthe signal as a result of the electrical characteristic of the sensor.

FIG. 8 is an example of a sensor graph that plots an electricalcharacteristic versus a condition. The sensor has a substantially linearregion in which an incremental change in a condition produces acorresponding incremental change in the electrical characteristic. Thegraph shows two types of electrical characteristics: one that increasesas the condition increases and the other that decreases and thecondition increases. As an example of the first type, impedance of atemperature sensor increases and the temperature increases. As anexample of a second type, a capacitance touch sensor decreases incapacitance as a touch is sensed.

FIG. 9 is a schematic block diagram of another example of a power signalgraph in which the electrical characteristic or change in electricalcharacteristic of the sensor is affecting the power signal. In thisexample, the effect of the electrical characteristic or change inelectrical characteristic of the sensor reduced the DC component but hadlittle to no effect on the oscillating component. For example, theelectrical characteristic is resistance. In this example, the resistanceor change in resistance of the sensor decreased the power signal,inferring an increase in resistance for a relatively constant current.

FIG. 10 is a schematic block diagram of another example of a powersignal graph in which the electrical characteristic or change inelectrical characteristic of the sensor is affecting the power signal.In this example, the effect of the electrical characteristic or changein electrical characteristic of the sensor reduced magnitude of theoscillating component but had little to no effect on the DC component.For example, the electrical characteristic is impedance of a capacitorand/or an inductor. In this example, the impedance or change inimpedance of the sensor decreased the magnitude of the oscillatingsignal component, inferring an increase in impedance for a relativelyconstant current.

FIG. 11 is a schematic block diagram of another example of a powersignal graph in which the electrical characteristic or change inelectrical characteristic of the sensor is affecting the power signal.In this example, the effect of the electrical characteristic or changein electrical characteristic of the sensor shifted frequency of theoscillating component but had little to no effect on the DC component.For example, the electrical characteristic is reactance of a capacitorand/or an inductor. In this example, the reactance or change inreactance of the sensor shifted frequency of the oscillating signalcomponent, inferring an increase in reactance (e.g., sensor isfunctioning as an integrator or phase shift circuit).

FIG. 11A is a schematic block diagram of another example of a powersignal graph in which the electrical characteristic or change inelectrical characteristic of the sensor is affecting the power signal.In this example, the effect of the electrical characteristic or changein electrical characteristic of the sensor changes the frequency of theoscillating component but had little to no effect on the DC component.For example, the sensor includes two transducers that oscillate atdifferent frequencies. The first transducer receives the power signal ata frequency of f1 and converts it into a first physical condition. Thesecond transducer is stimulated by the first physical condition tocreate an electrical signal at a different frequency f2. In thisexample, the first and second transducers of the sensor change thefrequency of the oscillating signal component, which allows for moregranular sensing and/or a broader range of sensing.

FIG. 12 is a schematic block diagram of an embodiment of a power signalchange detection circuit 112 receiving the affected power signal 118 andthe power signal 116 as generated to produce, therefrom, the signalrepresentative 120 of the power signal change. The affect 118 on thepower signal is the result of an electrical characteristic and/or changein the electrical characteristic of a sensor; a few examples of theaffects are shown in FIGS. 8-11A.

In an embodiment, the power signal change detection circuit 112 detect achange in the DC component 122 and/or the oscillating component 124 ofthe power signal 116. The power signal change detection circuit 112 thengenerates the signal representative 120 of the change to the powersignal based on the change to the power signal. For example, the changeto the power signal results from the impedance of the sensor and/or achange in impedance of the sensor. The representative signal 120 isreflective of the change in the power signal and/or in the change in thesensor's impedance.

In an embodiment, the power signal change detection circuit 112 isoperable to detect a change to the oscillating component at a frequency,which may be a phase shift, frequency change, and/or change in magnitudeof the oscillating component. The power signal change detection circuit112 is also operable to generate the signal representative of the changeto the power signal based on the change to the oscillating component atthe frequency. The power signal change detection circuit 112 is furtheroperable to provide feedback to the power source circuit 110 regardingthe oscillating component. The feedback allows the power source circuit110 to regulate the oscillating component at the desired frequency,phase, and/or magnitude.

FIG. 13 is a schematic block diagram of another embodiment of a drivesense circuit 28-b includes a change detection circuit 150, a regulationcircuit 152, and a power source circuit 154. The drive-sense circuit28-b is coupled to the sensor 30, which includes a transducer that hasvarying electrical characteristics (e.g., capacitance, inductance,impedance, current, voltage, etc.) based on varying physical conditions114 (e.g., pressure, temperature, biological, chemical, etc.).

The power source circuit 154 is operably coupled to the sensor 30 and,when enabled (e.g., from a control signal from the processing module 42,power is applied, a switch is closed, a reference signal is received,etc.) provides a power signal 158 to the sensor 30. The power sourcecircuit 154 may be a voltage supply circuit (e.g., a battery, a linearregulator, an unregulated DC-to-DC converter, etc.) to produce avoltage-based power signal or a current supply circuit (e.g., a currentsource circuit, a current mirror circuit, etc.) to produce acurrent-based power signal. The power source circuit 154 generates thepower signal 158 to include a DC (direct current) component and anoscillating component.

When receiving the power signal 158 and when exposed to a condition 114,an electrical characteristic of the sensor affects 160 the power signal.When the change detection circuit 150 is enabled, it detects the affect160 on the power signal as a result of the electrical characteristic ofthe sensor 30. The change detection circuit 150 is further operable togenerate a signal 120 that is representative of change to the powersignal based on the detected effect on the power signal.

The regulation circuit 152, when its enabled, generates regulationsignal 156 to regulate the DC component to a desired DC level and/orregulate the oscillating component to a desired oscillating level (e.g.,magnitude, phase, and/or frequency) based on the signal 120 that isrepresentative of the change to the power signal. The power sourcecircuit 154 utilizes the regulation signal 156 to keep the power signalat a desired setting 158 regardless of the electrical characteristic ofthe sensor. In this manner, the amount of regulation is indicative ofthe affect the electrical characteristic had on the power signal.

In an example, the power source circuit 158 is a DC-DC converteroperable to provide a regulated power signal having DC and ACcomponents. The change detection circuit 150 is a comparator and theregulation circuit 152 is a pulse width modulator to produce theregulation signal 156. The comparator compares the power signal 158,which is affected by the sensor, with a reference signal that includesDC and AC components. When the electrical characteristics is at a firstlevel (e.g., a first impedance), the power signal is regulated toprovide a voltage and current such that the power signal substantiallyresembles the reference signal.

When the electrical characteristics changes to a second level (e.g., asecond impedance), the change detection circuit 150 detects a change inthe DC and/or AC component of the power signal 158 and generates therepresentative signal 120, which indicates the changes. The regulationcircuit 152 detects the change in the representative signal 120 andcreates the regulation signal to substantially remove the effect on thepower signal. The regulation of the power signal 158 may be done byregulating the magnitude of the DC and/or AC components, by adjustingthe frequency of AC component, and/or by adjusting the phase of the ACcomponent.

It is noted that terminologies as may be used herein such as bit stream,stream, signal sequence, etc. (or their equivalents) have been usedinterchangeably to describe digital information whose contentcorresponds to any of a number of desired types (e.g., data, video,speech, text, graphics, audio, etc. any of which may generally bereferred to as ‘data’).

FIG. 14 is a schematic block diagram of an embodiment for providing areference signal waveform for a drive-sense circuit. In an example, asinusoidal waveform, such as oscillating component 124 is generated byreference signal generator 149, which is coupled to change detectioncircuit 150. Reference signal generator 149 can be a phase-locked loop(PLL) a crystal oscillator, a digital frequency synthesizer, and/or anyother signal source that can provide a sinusoidal signal of desiredfrequency, phase shift, and/or magnitude.

In general, a power source circuit 154 produces a source signal 158 thatis regulated to substantially match the sinusoidal reference signal 157.For example, the sinusoidal signal generated by reference signalgenerator 149 is useful when sensor 30 is one of a plurality of sensorssensing capacitance changes of a touch screen display. In such anenvironment, the use of a sinusoidal reference signal is readilygenerating and also does not introduce harmonics that may adverselyaffect the operation of the drive sense circuit, the touch screenoperation of the display, and/or the display operation of the display.

The output of power source circuit 154 (source signal 158) and referencesignal generator output (such as reference signal 157) are coupled tothe inputs of Op-amp 151, the output of which is coupled to analog todigital converter (ADC) 212. Signal 120, which represents the sourcesignal change is output by ADC 212 which output is also input toregulation circuit 152 and converted by digital to analog converter(DAC) 214; the output of regulation circuit 152 is coupled to powersource circuit 154 to provide regulation signal 156 to power sourcecircuit 154. The sinusoidal signal generated by reference signalgenerator 149 is non-linear signal and therefore has non-linearresolution.

FIG. 15 is a schematic block diagram of an embodiment of a data sensingcircuit 200 that includes an analog time domain circuit 202, an analogto digital circuit 204, and a digital frequency domain circuit 206. Thedata sensing circuit 200 operates in the time domain on information thatis in the frequency domain. In general, the data sensing circuit 200generates digital data 216 based on an analog frequency domain signal210 (e.g., an analog signal with data in the frequency domain) and areference signal 208. The resulting digital data 216 may be the desiredoutput data or may require further processing to obtain the desired dataoutput.

In an example of operation, the analog time domain circuit 202 outputs asignal component of the analog frequency domain signal 210 to a device218. The analog time domain circuit 202 includes a regulated sourcecircuit to generate the signal component. In one embodiment, theregulated source circuit is a dependent current source that is regulatedto a specific current value based on the reference signal 208. Inanother embodiment, the regulated source circuit is a voltage circuit(e.g., a linear regulator, a DC-DC converter, a battery, etc.) thatgenerates a regulated voltage based on the reference signal 208.

The device 218 alters the signal component to produce the analogfrequency domain signal 210, where the altering of the signal componentat a particular rate to represent input data. The inverse of the datarate corresponds to the frequency of the analog frequency domain signal210; thus, the signal in the analog domain and the data is in thefrequency domain. As an example, the signal component produced by theanalog time domain circuit 202 is a DC voltage (e.g., 0.25 volts to 5volts or more), which corresponds to the reference signal 208. Thedevice 218 alters the signal component by varying the loading on thesignal component to affect the voltage and/or current of the signalcomponent thereby created the analog frequency domain signal 210 (e.g.,the signal component plus the effects of altering).

As a specific example, the device 218 changes its resistance at aparticular rate (e.g., 10 Hz to 100 MHz or more) to represent the inputdata. An increase in resistance decreases voltage for a constantcurrent, decreases current for a constant voltage, or decreases bothvoltage and current of the signal component. A decrease in resistanceincreases the voltage for a constant current, increases the current fora constant voltage, or increases both voltage and current of the signalcomponent. The increasing and decreasing of the resistance of the deviceat the particular rate is representative of the input data. The numberof different resistance levels corresponds to the data level, where Nequals the number of unique data values per cycle of the data rate,where N is an integer of 2 or more. For instance, when N=2, there aretwo data levels (e.g., a logic “0” for a first resistance and a logic“1” for a second resistance) and when N=10, there are ten data levels(e.g., 0 through 9).

As another example of producing the analog frequency domain signal 210,the signal component produced by the analog time domain circuit 202includes an oscillating component (e.g., a sine wave, a triangular wave,square wave, saw-tooth wave, etc. with a peak to peak voltage of a fewmillivolts to 5 volts or more having a frequency of a 100 Hz to 1 MHz ormore), which corresponds to the reference signal 208. In this example,the device changes its impedance (e.g., capacitance, inductance, and/orresistance) at a particular rate (e.g., fx of 10 Hz to 100 MHz or more)to represent the input data. An increase in impedance decreases voltagefor a constant current, decreases current for a constant voltage, ordecreases both voltage and current of the signal component. A decreasein impedance increases the voltage for a constant current, increases thecurrent for a constant voltage, or increases both voltage and current ofthe signal component. The increasing and decreasing of the impedance ofthe device at the particular rate is representative of the input data.

Continuing with the example of operation, the analog time domain circuit202 uses the reference signal 208 in comparison to the analog frequencydomain signal 210 to create an analog frequency domain error correctionsignal 212. The analog frequency domain error correction signal 212 isrepresentative of the error correction needed to keep the signalcomponent and hence the analog frequency domain signal 210 substantiallymatching the reference signal. The error correction is representative ofthe frequency domain data that is embedded in the altering of the signalcomponent.

The analog to digital circuit 204 (e.g., an “n”-bit analog to digitalconverter, where n is an integer equal to or greater than 1) convertsthe analog frequency domain error correction signal 212 into a digitalfrequency domain error correction signal 214. The error correction,which is representative of the frequency domain data, is substantiallypreserved in the digital domain.

The digital frequency domain circuit 206 operates in the frequencydomain to recover the digital data 216. For example, the digitalfrequency domain circuit 206 includes one or more finite impulseresponse (FIR) filters, one or more cascaded integrated comb (CIC)filters, one or more infinite impulse response (FIR) filters, one ormore decimation stages, one or more fast Fourier transform (FFT)filters, and/or one or more discrete Fourier transform (DFT) filters.

FIG. 16 is a schematic block diagram of another embodiment of a datasensing circuit 200 that includes an analog time domain circuit 202-1,an analog to digital circuit 204, a digital frequency domain circuit206; and a digital to analog feedback circuit 220. This data sensingcircuit 200 operates similarly to the data sensing circuit 200 of FIG.10 with the following differences. The feedback for regulating thesignal component via the regulated source circuit within the analog timedomain circuit 201-1 is from the digital to analog feedback circuit 220(e.g., an “n”-bit digital to analog converter, when n is an integerequal to or greater than 1).

FIG. 17 is a schematic block diagram of another embodiment of a datacircuit 230 that includes a drive sense circuit 28, a plurality ofdigital bandpass filters (BPF) circuits 232-236, and a plurality of datasources (1 through n). The drive sense circuit 28 produces a drivesignal component of a drive & sense signal 238 (e.g., the drive part ofsignal 238) based on the reference signal 208 as previously discussed.The data sources operate at different frequencies to embed frequencydomain data into the drive & sense signal 238 (e.g., the sense part ofsignal 238). Each of the data sources operates similarly to the device218 of FIG. 10 to embed the data into the signal 238 by varying theloading on the drive component of signal 238.

In an example of operation, data source 1 alters the drive signalcomponent of the drive & sense signal 238 at a first frequency f1; datasource 2 alters the drive signal component of the drive & sense signal238 at a second frequency f2; and data source n alters the drive signalcomponent of the drive & sense signal 238 at an “nth” frequency fn. Thedrive sense circuit 28 regulates the drive & sense signal 238 tosubstantially match the reference signal 208, which may be similar toreference signal 157 of FIG. 14.

The drive sense circuit 28 outputs a signal 120 that is representativeof changes to the drive & sense signal 238 based on the regulation ofthe drive & sense signal 238. Each of the digital BPF circuits 232receives the signal 120 and is tuned to extract data therefromcorresponding to one of the data sources. For example, digital BPFcircuit 232 is tuned to extract the data at frequency f1 of the datasource 1 to produce one or more digital values representing the firstdata 240. The second digital BPF circuit 234 is tuned to extract thedata at frequency f2 of the data source 2 to produce one or more digitalvalues representing the second data 242. The nth digital BPF circuit 236is tuned to extract the data at frequency fn of the data source n toproduce one or more digital values representing the nth data 244. Eachof the digital BPF circuits 232-236 includes one or more finite impulseresponse (FIR) filters, one or more cascaded integrated comb (CIC)filters, one or more infinite impulse response (IIR) filters, one ormore decimation stages, one or more fast Fourier transform (FFT)filters, and/or one or more discrete Fourier transform (DFT) filters.

FIG. 18 is a schematic block diagram of an embodiment of an analog todigital conversion circuit 246 that includes an analog to digitalconverter (ADC) 258, a digital decimation filtering circuit 248, adigital bandpass filter (BPF) circuit 250, and a processing module 252.The ADC 258 may be implemented in a variety of ways. For example, theADC 258 is the ADC converter 212 of drive sense circuit 28 of previousFigures. As another example, the ADC 258 is implemented as a flash ADC,a successive approximation ADC, a ramp-compare ADC, a Wilkinson ADC, anintegrating ADC, and/or a delta encoded ADC. As yet another example, theADC 258 is implemented as a sigma-delta ADC.

The digital decimation filtering circuit 248 includes one or more finiteimpulse response (FIR) filters, one or more cascaded integrated comb(CIC) filters, one or more infinite impulse response (FIR) filters, oneor more decimation stages, one or more fast Fourier transform (FFT)filters, and/or one or more discrete Fourier transform (DFT) filters,one or more polyphase filters, and one or more decimation stages. Thedigital bandpass filter (BPF) circuit 250 includes one or more finiteimpulse response (FIR) filters, one or more cascaded integrated comb(CIC) filters, one or more infinite impulse response (FIR) filters, oneor more decimation stages, one or more fast Fourier transform (FFT)filters, and/or one or more discrete Fourier transform (DFT) filters,and one or more polyphase filters. BPF 250 includes a plurality of tapshaving coefficients set to produce a bandpass region approximatelycentered at the oscillation frequency of the analog input signal andhaving a bandwidth tuned for filtering a pure tone (e.g., s1).

Sampling frequencies of the stages of the analog to digital conversioncircuit 246 are set as multiples of the data output rate. For example,the data output 256 rate is 300 Hz thus sampling frequencies aremultiples of 300 Hz. For example, ADC circuit 258 oversamples the analoginput signal at a sampling frequency (fs) of 2¹⁷*300 Hz (approximately39.32 MHz). The analog input signal is said to be oversampled when thesampling frequency is more than the Nyquist sampling frequency (e.g., 40KHz-400 KHz when the oscillating frequency is 20 KHz-200 KHz). Settingthe sampling frequency at a frequency much higher than the Nyquistsampling frequency results in a significantly oversampled analog signal.Oversampling of the analog signal allows for narrower bandpass filteringand improves signal to noise ratio (SNR).

In an example of operation, the ADC 258 converts an analog signal thatincludes a set of pure tone components (e.g., one or more pure tonecomponents, each having an oscillation frequency) into a digital signalof the one or more pure tone components. For example, an input analogsignal has a pure tone (e.g., a sinusoidal signal, a DC signal, arepetitive signal, and/or a combination thereof) having a DC componentand/or an oscillation frequency at f1 (e.g., a frequency in the audiorange, in the range 20 KHz-200 KHz, or more). As a specific example, theADC is a sigma-delta ADC that oversamples the analog input signal atclock rate of approximately 39.32 MHz (e.g., 300*2¹⁷) and, as such,pushes low frequency noise up to higher frequencies outside the band ofinterest. An example of a sigma-delta ADC will be discussed in greaterdetail with reference to FIG. 23.

Continuing with the specific example, the ADC 258 produces a 1-bitdigital output at approximately 39.32 MHz representative of the analogsignal. In an embodiment, the analog signal includes an error correctionsignal s1 at frequency f1, which represents the frequency domain dataembedded in the analog input signal and is substantially preserved inthe digital domain (e.g., as discussed in FIGS. 14-15).

The digital decimation filtering circuit 248 takes the output from ADCcircuit 258 (e.g., 1-bit digital output at approximately 39.32 MHz) andconverts it to another digital signal having another data rate frequencythat is a multiple of the data output rate (e.g., 300 Hz). In thisexample, digital decimation filtering circuit 248 has an output rate(fd) of 2¹²*300 HZ (approximately 1.23 MHz).

As a more specific example, the digital decimation filtering circuit 248converts the 1-bit digital output at approximately 39.32 MHz into an18-bit output at 2¹²*300 HZ (approximately 1.23 MHz) representing errorcorrection signal s1 at frequency f1. The ratio between the samplingrate (fs) and the digital decimation filtering circuit 248's output rate(fd) (e.g., fs/fd) is equal to the number of ADC 258 samples per outputof digital decimation filtering circuit 248. For example, 39.32 MHz/1.23MHz=32. Therefore, digital decimation filtering circuit 248 has adecimation rate of 32. In the time is takes ADC 258 to output 32 1-bitsamples, 1 18-bit output is produced by digital decimation filteringcircuit 248. The digital decimation filtering circuit 248 will bediscussed in greater detail with reference to FIGS. 31-41.

The digital BPF circuit 250 takes the output of the digital decimationfiltering circuit 248 (e.g., the 18-bit output at approximately 1.23MHz) and bandpass filters it. The digital BPF circuit 250 applies anarrow bandpass filter, centered at f1, and outputs an affect value 254having real and imaginary components. Because the data (e.g., errorcorrection signal) is embedded in a sinusoid (e.g., a pure tone) thedesired information is at frequency f1 and is based on magnitude and/orphase. Therefore, the bandpass filter can be very narrow (e.g., 1% to20% of channel spacing and, as a specific example about 5% the channelspacing 255 (e.g., for a channel spacing of 300 Hz, a 10 Hz bandpassfilter may be used)) to capture the desired signal. In an embodiment,the digital BPF circuit 250 has a tap-length of 4096 (e.g., in the timeit takes digital decimation filtering circuit 248 to output 4096 18-bitoutputs at approximately 1.23 MHz, digital BPF circuit outputs 1 48-bitaffect value at the output rate of 300 Hz). The digital BPF circuit 250will be discussed in greater detail with reference to FIGS. 42-53.

Processing module 252 interprets the imaginary and real components ofthe affect value 254 to produce data output 256. Affect value 254 is avector (i.e., a phasor complex number) having a real component and animaginary component representing a sinusoidal function that has a peakmagnitude (i.e., amplitude) and direction (i.e., phase). For example,affect value 254 is one 48-bit value having a 24-bit real component anda 24-bit imaginary component. In the complex domain, voltages andcurrents are phasors and resistances, capacitances, and inductances arereplaced with complex impedances (e.g., ZR=R, ZL=jfL, andZC=1/(jfC)=−j/(fC)). Since voltage (V)=current (I)*impedance (Z), theprocessing module 252 determines a capacitance or other impedance valuefrom voltage and current vectors of the affect value 254 (e.g., adecrease in impedance increases the voltage for a constant current,increases the current for a constant voltage, or increases both voltageand current of the signal component). The increasing and decreasing ofimpedance at a particular rate is representative of the input data. Theimpedance value or change in impedance value determined is output asdata output 256 at the example output rate of 300 Hz.

FIG. 19 is a schematic block diagram of another embodiment of analog todigital conversion circuit 246 that includes analog to digital converter(ADC) 258, digital decimation filtering circuit 248, a plurality ofdigital bandpass filter (BPF) circuits 250, and processing module 252.Analog to digital conversion circuit 246 of FIG. 19 operates similarlyto the example of FIG. 18 except a plurality of digital BPF circuits areincluded for filtering a plurality of pure tones.

In an example of operation, the ADC 258 converts an analog signal havinga set of pure tone components (e.g., signals s1-sn) into a set ofdigital signals (s1-sn) at the oscillation frequencies (e.g., f1-fn).For example, a first tone of the input analog signal has an oscillationfrequency of f1 (e.g., 100 KHz), which, for example, is used for a firstself-capacitance measurement on a touch screen display, a second tone ofthe input analog signal has an oscillation frequency of f2 (e.g., 100.3KHz), which, for example, is used for a first mutual-capacitancemeasurement on a touch screen display, and an nth tone of the inputanalog signal has an oscillation frequency of fn (e.g., 100 KHz+300nHz), which, for example, is for an nth mutual-capacitance measurementon a touch screen display. Frequencies f1-fn span n channels and areequally separated by a channel spacing 255. For example, channel spacing255 is equal to output data rate of 300 Hz.

The digital decimation filtering circuit 248 takes the output from theADC 258 (e.g., via a n-line parallel bus) and converts the signals toother digital signals having another data rate frequency that is amultiple of the data output rate (e.g., 300 Hz). In this example,digital decimation filtering circuit 248 has an output rate (fd) of2¹²*300 HZ (approximately 1.23 MHz). For example, the digital decimationfiltering circuit 248 converts the 1-bit ADC output at approximately39.32 MHz representing digital signals s1-sn at frequencies f1-fn to an18-bit output at 2¹²*300 HZ (approximately 1.23 MHz) representingsignals s1-sn at frequencies f1-fn.

Each of the digital BPF circuits 1-n 250 includes a plurality of tapshaving coefficients set to produce a bandpass region approximatelycentered at the oscillation frequency of the analog input signal andhaving a bandwidth tuned for filtering a pure tone. For example, digitalBPF circuit 1 250 has a bandwidth tuned for filtering f1, digital BPFcircuit 2 250 has a bandwidth tuned for filtering f2, and digital BPFcircuit n 250 has a bandwidth tuned for filtering fn. Digital BPFcircuits 1-n 250 take the output from the from the digital decimationfiltering circuit 248 (e.g., n 18-bit outputs at approximately 1.23 MHzwith error correction signals s1-sn at frequencies f1-fn via a bus) andshifts each signal to the bandpass for a frequencies f1-fn.

Digital BPF circuits 1-n 250 each apply a very narrow bandpass filterand output a corresponding affect value 1-n 254 having real andimaginary components. Because data is embedding in each sinusoid signal(s1-sn) (e.g., a pure tone) the desired information is at frequenciesf1-fn and based on magnitude and/or phase. Therefore, the bandpassfilters can be very narrow (e.g., less than 0.05 the channel spacing(e.g., 10 Hz)) to capture the desired signals.

Processing module 252 interprets the imaginary and real components ofthe affect values 1-n 254 to produce data outputs 1-n 256. Affect values1-n 254 are vectors (i.e., a phasor complex numbers) each having a realcomponent and an imaginary component representing a sinusoidal functionthat has a peak magnitude (i.e., amplitude) and direction (i.e., phase).For example, an affect value is one 48-bit value having a 24-bit realcomponent and a 24-bit imaginary component. In the complex domain,voltages and currents are phasors and resistances, capacitances, andinductances are replaced with complex impedances (e.g., ZR=R, ZL=jfL,and ZC=1/(jfC)=−j/(fC)). Since voltage (V)=current (I)*impedance (Z),the processing module 252 determines capacitance or other impedancevalues from voltage and/or current vectors represented by affect values1-n 254. The impedance values or changes in impedance values determinedare output as data outputs 1-n 256. Data output 256 is output separatelyor in parallel at the output data rate (e.g., 300 Hz).

FIGS. 20A-20B are example graphs that plot condition verses capacitance(e.g., of an electrode of a touch screen display). In a touch screendisplay example, an electrode has a self-capacitance and mutualcapacitance. A finger capacitance or a pen capacitance (e.g., a touch)raises self-capacitance of electrodes which decreases the impedance fora given frequency. As shown in FIG. 20A, the mutual capacitancedecreases with a touch and the self-capacitance and pen-capacitanceincreases with a touch. As shown in FIG. 20B, the mutual capacitance,pen-capacitance, and self-capacitance for a no-touch condition are shownto be about the same magnitude but are different than when under a touchcondition. For instance, the mutual capacitance decreases as a result ofa touch, while self-capacitance and pen-capacitance each increases as aresult of a touch.

FIG. 21 is an example graph that plots impedance verses frequency for aninput that has a primarily capacitive load. Being based on capacitance(self, pen, and/or mutual), as the frequency increases for a fixedcapacitance, the impedance decreases based on ½πfC, where f is thefrequency and C is the capacitance.

FIG. 22 is an example of affect values 254-1 and 254-2. When the DCcomponent embedded in the analog input signal represents a voltage at aconstant current, an affect value represents a voltage vector having animaginary component and a real component. The processing module 252determines capacitance changes (e.g., self, pen, mutual, etc.) fromvoltage vectors (e.g., impedance (Z)=voltage (V)/current (I) andZC=1/(jfC)=−j/(fC)) and interprets whether the change represents a touchor no touch condition.

FIG. 23 is a schematic block diagram of an embodiment of a sigma deltaanalog to digital (ADC) circuit. Sigma delta (ADC) circuit 258 is anexample of ADC 258 of FIGS. 18 and 19 and includes oversamplingmodulator 260 and digital decimation filtering circuit 248. In anexample of operation, the ADC circuit 258 converts an analog inputsignal 262 having an oscillation frequency and a set of pure tonecomponents into an 18-bit output at a rate of approximately 1.23 MHz.For example, an input analog signal 262 has an oscillation frequency offi (e.g., 20 KHz-200 KHz) and a pure tone component s1.

In this example, oversampling modulator 260 is a 1-bit ADC sigma-deltamodulator. Oversampling modulator 260 oversamples the analog inputsignal 262 at a sampling frequency (fs) of 2¹⁷*300 Hz (approximately39.32 MHz) in this example. Oversampling modulator 260 produces a 1-bitADC output at 39.32 MHz representing error correction signal s1 embeddedin the sinusoidal signal at frequency f1. Error correction signal s1 isrepresentative of the frequency domain data embedded in the analog inputsignal and is substantially preserved in the digital domain.

Digital decimation filtering circuit 248 includes one or more finiteimpulse response (FIR) filters, one or more cascaded integrated comb(CIC) filters, one or more infinite impulse response (FIR) filters, oneor more decimation stages, one or more fast Fourier transform (FFT)filters, and/or one or more discrete Fourier transform (DFT) filters,one or more polyphase filters, and one or more decimation stages.Digital decimation filtering circuit 248 takes the output fromoversampling modulator 260 (e.g., 1-bit ADC output at approximately39.32 MHz representing error correction signal s1 at frequency f1) andfilters and down converts it to another digital signal having anotherdata rate frequency. In this example, digital decimation filteringcircuit 248 has an output rate (fd) of 2¹²*300 HZ (approximately 1.23MHz).

For example, the digital decimation filtering circuit 248 converts the1-bit ADC output at approximately 39.32 MHz representing errorcorrection signal s1 at frequency f1 to an 18-bit output at 2¹²*300 HZ(approximately 1.23 MHz) representing error correction signal s1 atfrequency f1. The ratio between the sampling rate (fs) and the digitaldecimation filtering circuit 248's output rate (fd) (e.g., fs/fd) isequal to the number of samples taken by the oversampling modulator 260per output of the digital decimation filtering circuit 248. For example,39.32 MHz/1.23 MHz=32. Therefore, digital decimation filtering circuit248 has a decimation rate of 32.

FIG. 24 is an example of quantization noise of a sigma deltaoversampling modulator 260 of FIG. 23. Sigma-delta ADCs implement noiseshaping (i.e., a function that effectively pushes low frequency noise upto higher frequencies outside the band of interest) making it suitablefor high precision, high resolution applications. Oversampling modulator260 of FIG. 23 moves quantization noise 264 to higher frequencies. Theorder of the sigma delta oversampling modulator varies the noiseshaping.

As shown, quantization noise 264 starts low at zero Hz, rises and thenlevels off at the oversampling modulator's sampling frequency (fs).Multi-order sigma delta modulators shape the quantization noise 264 tohigher frequencies than lower-order sigma delta modulators. For example,the third-order sigma delta modulator example shows much more noise nearfrequency fs in comparison to the first-order sigma delta modulator butnoise near lower frequencies is much less. The output of digitaldecimation filtering circuit 248 of FIG. 23 includes frequencies from 0to frequency fd and thus a good portion of the quantization noise 264exists in the output of all three examples. However, very narrowbandpass filtering (e.g., by digital BPF circuit 250 as discussed inprevious Figures) isolates the signals of interest at the lowerfrequencies such that noise near fd is also removed.

FIG. 25 is a schematic block diagram of example outputs of the differentstages of the analog to digital conversion circuit 246 of FIGS. 18 and19. In this example, analog to digital (ADC) circuit 258 produces a1-bit ADC output at 2¹⁷*300 Hz (approximately 39.32 MHz). Therefore,there are 2¹⁷ (or 131,072) 1-bit samples of the analog input signal perdata output clock cycle (e.g., 300 Hz in this example). Digitaldecimation filtering circuit 248 produces an 18-bit output at 2¹²*300 Hz(approximately 1.23 MHz). 2¹⁷/2¹² is equal to 2⁵ or 32; therefore, inthe time the ADC circuit 258 outputs 32 1-bit samples and the digitaldecimation filtering circuit 248 is able to output one 18-bit value asshown.

BPF circuits 250 output one 48-bit affect value having a 24-bit realcomponent and a 24-bit imaginary component at the data output clock rateof 300 Hz. Therefore, there are 2¹² (or 4096) 18-bit values per dataoutput clock cycle (e.g., 300 Hz in this example). In other words, inthe time it takes digital decimation filtering circuit 248 to output4096 18-bit values, the one or more digital BPF circuits 250 output one48-bit affect value having a 24-bit real component and a 24-bitimaginary component at the data output clock rate of 300 Hz.

FIG. 26 is an example of sampling an analog signal 262 to produce adigitized signal 270. In this example, analog signal 262 is sampled at 8points per cycle (s0-s7) to create a digitized signal of 8 discretepoints representative of the analog signal 262.

FIG. 27 is a schematic block diagram of a digital filter implementing amultiply-accumulate function. The digital filter shown is designed with8 stages (e.g., taps) in order to capture the 8 discrete points of thedigitized signal of FIG. 26. When the 8 stages capture the points in thepattern shown in FIG. 26, the digital filter produces a filtered output272 (e.g., a pulse representative of an n-bit digital logic value). Theinput signal (e.g., digitized signal 270) enters the digital filter atstage 0 where it is multiplied by coefficient h0 and also input intostage 1. Stages 1-7 each include a unit delay Z⁻¹ in Z-transformnotation to provide delayed inputs (taps) to each stage's multiplicationoperation (i.e., the input signal is multiplied by the next coefficient(e.g., h1-h6) after a delay Z⁻¹). The results of the multiplicationoperation from each stage are added (i.e., accumulated) to create thefiltered output. The series of multiply accumulate functions is alsoreferred to as a moving average. The more taps the filter has, the morecomputationally extensive the output becomes.

FIG. 28 is a schematic block diagram of a digital filter implementing amultiply-accumulate function. The digital filter operates similarly tothe digital filter of FIG. 27 and is shown here for convenience.

FIG. 29 is an example of a digitized signal 270. At a point in time,digitized signal 270 has a particular pattern. For example, the patternshown is one cycle of a sinusoidal signal. Coefficients h0-h7 of thedigital filter of FIG. 28 can be set so that only something close to thedesired pattern produces a viable output.

FIG. 30 is an example of producing a digital filtered output 272. Asdigitized signal 270 of FIG. 29 moves through the stages of the digitalfilter (e.g., of FIGS. 27 and 28), coefficients h0-h7 at stages 0-7 areset to look for the pattern shown in FIG. 29 (i.e., the coefficients setthe center frequency of the bandpass filter, the bandwidth of thebandpass filter, and the roll-off of the bandpass filter). When thepattern shown in FIG. 29 (or something fairly close to the pattern) isrecognized, the bandpass filter produces an output indicating thepresence of the signal (e.g., a magnitude and/or phase of a sinusoidalsignal). As shown, at stage 7 and at time t7, the filter recognizes thatthe pattern shown in FIG. 29 has moved through stages 0-7 and thereforeproduces a filtered output 272 at time t7.

The filtered output 272 may be a pulse representative of an n-bitdigital logic value. For example, a digitized sinusoidal signal of afirst amplitude may produce pulse representative of a 1-bit digitallogic of zero and a digitized sinusoidal signal of a second amplitudemay produce pulse representative of a 1-bit digital logic of one.Therefore, digital data (e.g., signal s1 of FIG. 18-19) can be embeddedin an analog signal and extracted via digital filtering.

FIG. 31 is a schematic block diagram of an embodiment of a digitaldecimation filtering circuit 248. Digital decimation filtering circuit248 includes anti-aliasing filter 274 and decimator 276. In general,digital decimation filtering circuit 248 filters high frequencycomponents of the input signal and reduces the sampling rate so that thenext stage of analog to digital conversion circuit 246 can operate moreefficiently.

Digital decimation filtering circuit 248 receives a 1-bit ADC outputstream at approximately 39.32 MHz from ADC 258 or oversampling modulator260 when ADC 258 is sigma delta ADC 258 of FIG. 23. By oversampling theanalog input signal, quantization noise 264 is spread out over a widerbandwidth. When ADC 258 is a first order sigma delta ADC, the outputfrom the oversampling modulator 260 includes quantization noise 264 thatis noise shaped to be greatest at the sampling frequency (fs) of theoversampling modulator 260 (e.g., 39.32 MHz) as shown.

Anti-aliasing filter 274 is a lowpass filter averaging filter (e.g., oneor more finite impulse response (FIR) filters, one or more comb filters,one or more raised cosine filters, one or more cascaded integrated comb(CIC) filters, one or more infinite impulse response (IIR) filters, oneor more decimation stages, one or more fast Fourier transform (FFT)filters, and/or one or more discrete Fourier transform (DFT) filters,etc.) that samples the 1-bit ADC output and provides a cutoff frequencyto remove or attenuate signals (e.g., quantization noise 264) at higherfrequencies. Anti-aliasing filter 274 has a frequency response H(z).

Decimator 276 reduces the output rate of anti-aliasing filter 274 bythrowing away portions of anti-aliasing filter 274's output data. Inthis example, decimator 276 reduces the output rate of anti-aliasingfilter 274 (e.g., 39.32 MHz) by 32 to produce digital decimationfiltering circuit 248 output rate of 18-bit at approximately 1.23 MHz(e.g., 39.32 MHz/32=1.23 MHz). As shown, applying a low passanti-aliasing filter 274 with a cutoff frequency of fd and decimatingthe signal by 32 removes a portion of quantization noise between fd andfs.

FIG. 32 is an example frequency response H(z) of the anti-aliasingfilter 274. For example, anti-aliasing filter 274 is a finite impulseresponse (FIR) filter that cuts off frequencies higher than 1.23 MHz(i.e., the output rate of digital decimation filtering circuit 248). TheFIR filter has a sin x/x (e.g., or “sinc”) frequency response as shown.The sinc frequency response has a “notch” response (e.g., it can rejectthe line frequency when set to that frequency). The notch position isalso directly related to the output data rate. As shown, the first notchposition in FIG. 32 is located at the output rate of the digitaldecimation filtering circuit 248 output rate of approximately 1.23 MHz(e.g., the cutoff frequency). The sinc frequency response is equal tozero at integer multiples of the data rate (e.g., 2.46 MHz, 3.69 MHz,and so on). With a sampling rate of 39.32 MHz, the signal can containfrequency content up to 39.32 MHz/2=19.66 MHz according to Nyquistsampling theorem.

FIG. 33 is a schematic block diagram of an embodiment of anti-aliasingfilter 274. In this example, anti-aliasing filter 274 is implementing amultiply accumulate function as discussed in FIG. 27. For example,anti-aliasing filter 274 is a lowpass finite impulse response (FIR)filter having N number of taps. The number of taps selected inanti-aliasing filter 274 is related to the sampling frequency (e.g.,39.32 MHz), the desired cutoff or stopband frequency (e.g., 1.23 MHz),and several other desired filter properties. For example, increasing thenumber of taps in a FIR filter reduces noise, reduces transitionbandwidth between stopband and passband frequencies, and increasesattenuation in the stopband. However, the more taps a FIR filter has,the more computationally extensive it is (e.g., more multiplyaccumulates are required).

In a specific example, anti-aliasing filter 274 is a 128-tap FIR filter(e.g., the FIR filter has 128 frequency coefficients h0-h127) that cutsoff frequencies higher than 1.23 MHz (i.e., the output rate of digitaldecimation filtering circuit 248) and runs at the 1-bit ADC outputfrequency of 39.32 MHz. The 1-bit ADC output at of 39.32 MHz is a streamof 1-bit code in the time domain shown here as input signal x[n], wherex[n] includes n discrete points. The analog signal shown as a dottedline over the stream of 1-bit code shows a simplified example of how astream of 1-bit inputs can represent an analog signal. As discussed inprevious Figures, digital decimation filtering circuit 248 filters 32samples of input at a time. To accommodate for the 128-taps, the 32-bitinput can be padded with zeros.

The input signal enters the anti-aliasing filter 274 at stage 0 where itis multiplied by coefficient h0 and also input into stage 1. Stages1-127 each include a unit delay Z⁻¹ in Z-transform notation to providedelayed inputs (taps) to each stage's multiplication operation (i.e.,the input signal is multiplied by the next coefficient (e.g., h1-h127)after a delay Z⁻¹). The results of the multiplication operation fromeach stage are added (i.e., accumulated) to create the filtered output.The series of multiply accumulate functions is also referred to as amoving average. The more taps, the more computationally extensive theoutput becomes.

The output signal from the anti-aliasing filter 274 is equal toy[n]=Σ_(i=0) ^(N) h[i]·x[n−i] where N is 128 in this example. The outputequation is a summation of the convolution of the input signal with thefilter's coefficients. In the time domain, the 128-bit code trainresembles the original analog signal (here only 20 bits are shown forconvenience) and is responsible for high resolution. However, in thefrequency domain, anti-aliasing filter 274 only applies a low passfilter to the signal to attenuate the quantization noise. Therefore, theoutput signal is now a high-resolution digital version of the analoginput signal.

FIG. 34 is a schematic block diagram of an embodiment of a decimator276. Decimator 276 takes the output from the 128-tap anti-aliasingfilter 274 represented here as y[n]=y[0]+y[1]+ . . . +y[127] (note thatthe illustration only shows 20 samples for convenience) and throws outevery M calculation (e.g., where M is the decimation factor). Forexample, with a decimation factor of 32 and an input of 128 samples fromthe 128-tap anti-aliasing filter 274, decimator 276 outputs 4 outputsy[0]+y[1] (formerly y[31]), +y[2](formerly y[63])+y[3](formerly y[95]).From the summation of the four outputs, one 18-bit output is produced atthe output rate of approximately 1.23 MHz.

FIG. 35 is an example of a frequency band having frequency channels. Thefrequency band of interest 280 begins at f1 and ends at fn. Frequencyband of interest 280 includes channels 282 f1-fn spaced out at a desiredchannel spacing 255 (e.g., the data output rate of 300 Hz or anotherfrequency). As a specific example, frequency band of interest 280includes 128 channels where each channels contains a pure tone components1-s128 having frequencies f1-f128. With a channel spacing of 300 Hz(e.g., the data output rate), the frequency band of interest is 128×300Hz=38.4 KHz wide (i.e., n×channel spacing 200 Hz). If f1 is at 100 KHz,the frequency band of interest 280 spans from 100 Khz to 138.4 KHz.

FIG. 36 is a schematic block diagram of another embodiment of digitaldecimation filtering circuit 248. The digital decimation filteringcircuit 248 includes anti-aliasing filters 274-1 through 274-n anddecimators 276-1 through 276-n where n corresponds to n channels of1-bit ADC output. As an example, anti-aliasing filters 274-1 through274-n are 128-tap finite impulse response (FIR) filters. The n channelsof ADC output are delivered to digital decimation filtering circuit 248via an n-line parallel bus 284. Each channel of 1-bit data from the ADCis filtered with a corresponding anti-aliasing filter 274-1 through274-n and decimated by a factor of 32 by a corresponding decimator 276-1through 276-n to produce n outputs at the digital decimation filteringcircuit 248 output rate.

For example, digital decimation filtering circuit 248 takes 128 channelsof 1-bit output at 39.32 MHz from the ADC and filters each channel withan anti-aliasing filter with a decimation factor of 32 producing 12818-bit outputs at a sample rate of approximately 1.23 MHz. For example,the 128 18-bit outputs are multiplexed onto a single bus running atapproximately 157.29 MHz (i.e., 128 (2⁷) channels×the output rate 1.23MHz (2¹²×300 Hz) or 2¹⁹×300 Hz=approximately 157.29 MHz). As a specificexample, the output bus is a 16-bit bus with eight idle time slots(e.g., 8 bits are needed to multiplex the 128 channels which is an 8-bitbinary number). The output bus runs at 128 times the output rate toallow for each channel to run through each anti-aliasing filter 274-1through 274-n and be output onto a single bus. Alternatively, the 12818-bit outputs may be output in parallel.

FIG. 37 is a schematic block diagram of another embodiment of digitaldecimation filtering circuit 248. In contrast to the 128-tap finiteimpulse response (FIR) anti-aliasing filter 274 and decimator 276 ofFIGS. 31-36, the digital decimation filtering circuit 248 shown hereincludes 32 4-tap polyphase filters E₀(z)-E₃₁(z) with coefficientse(n)=h(32n+1), n=0 . . . 3 and l=0 . . . 31. Each polyphase filterincludes a delay (z⁻¹) and a decimator (↓32) producing a result that isadded by a summation network 278 to compute the final output.

In the example of FIGS. 31-36, the filter response is convolved with thefull signal and many points that were just calculated are thrown away(e.g., the signal is filtered then decimated). Polyphase filters are amore efficient implementation for digital decimation filtering circuit248 because the signal can be decimated prior to filtering andcalculations are not wasted. Further, each polyphase filter in thedigital decimation filtering circuit runs at the slower digitaldecimation filtering circuit 248 output rate of 1.23 MHz (in comparisonto the 128 FIR filter which runs at 39.32 MHz).

In this example, 32 polyphase filters are needed because the decimationrate is 32. Each sample on the input to digital decimation filteringcircuit 248 is delivered to just one of the polyphase filters. 32 1-bitinput samples (e.g., from the 1-bit ADC output stream at 39.32 MHz) areloaded into the 32 polyphase filters starting from the bottom (at stage0) and working up. After 32 1-bit samples are loaded, the polyphasefilters run to generate a single output point (e.g., an 18-bit output at1.23 MHz). The procedure is repeated for the next 32 samples.

FIG. 38 is a schematic block diagram of an example of polyphase filtersof digital decimation filtering circuit 248 shown in FIG. 37. Eachpolyphase filter E₀(z)-E₃₁(z) includes 4 coefficients (e.g., 4 taps).The frequency response of the 128-tap FIR filter discussed in previousFigures can be rewritten as a summation of the frequency response ofeach filter E₀(z)-E₃₁(z). Based on the decimation factor, the taps thatproduce an output can be included in one filter (e.g., E₀(z) includestaps h[0], h[32]z⁻¹, h[64]z⁻² and h[96]z⁻³ which extract data from theinput signal at every 32^(nd) point. The input signal x[n] can then bebroken up in order to decimate the signal prior to input into filtersE₀(z)-E₃₁(z). For example, x[n] values x[0], x[32], x[64], and x[96] areinput into filter E₀(z) which with produce the values needed fordecimation. Other inputs are multiplied by zero in order to not wastecalculations done by the filters. Summation network 278 adds the results(e.g., y[0], y[1], y[2], and y[3]) from the filters E₀(z)-E₃₁(z) toproduce an 18-bit output at the output rate of 1.23 MHz.

FIG. 39 is a schematic block diagram of another embodiment of thedigital decimation filtering circuit 248. In this example, digitaldecimation filtering circuit 248 filters and decimates (by a factor of32) 128 channels of 1-bit ADC output at 39.32 MHz through the structureshown. In contrast to the example shown in FIG. 36, where n (e.g., 128)separate anti-aliasing filters and decimators are required for eachinput channel, here each channel is processed by the same filterstructure. In order to filter and decimate 128 channels through onestructure, digital decimation filtering circuit 248 includes 32 shiftregister memories 286, 32 bit shifts 288, 32 4-tap polyphase filtersE0-E31 (e.g., of FIG. 37) implemented as 32 look-up tables (LUTs) 290,and summation tree 292.

Each shift register memory 286 contains a 5-bit register for each of the128 channels. In each 5-bit register, one of the bits is reserved fornew input while the other 4 contain the previous 4 binary inputs. Memoryis written to each shift memory registry 286 a column at a time at thesame frequency as the output sample rate (e.g., 1.23 Mz) however, memoryis read out a row at a time into the filter structure at a rate of 128times the output sample rate (e.g., 128×1.23 Mz=157.29 MHz). As shown,by the time 128 bits are input to each shift memory registry 286 at theoutput rate, 4 bits are input into each LUT 290 per cycle. With adecimation factor of 32, 4 bits are output for every 128 bits in thisexample. The structure of the shift memory registry 286 will bediscussed in greater detail with reference to FIG. 40.

The bit shift 288 removes the input bit and rearranges the other 4 bitsinto the correct address lines of the look up tables (LUTs) 290.Polyphase filters E0-E31 are implemented as a set of look-up tables(LUTs) 290. LUTs 290 store pre-computed product values corresponding topossible input values. Pre-computed product values are stored at amemory location whose address location is the same as the binary valueof the input value the product value corresponds to. For example, foreach polyphase filter E0-E31, the output for the 16 possiblecombinations of the 4 binary input taps are precomputed and stored inthe table. Each LUT takes the 4-bit input from the bit shift 288 anddetermines a precomputed value based on the address. The output fromeach LUT is 16-bits representative of 4 bits of input data (e.g., 1-bitper tap (4 taps), per input (4)). The 16-bits representative of 4 bitsof input data are put through the summation tree 292 to calculate the18-bit final result.

Digital decimation filtering circuit 248 runs at a rate of 128 times theoutput rate (e.g., 128×1.23 MHz=157.29 MHz) so that all 128 channels canbe processed through the same structure and all output data can beprocessed on the same output bus. Digital decimation filtering circuit248 processes 4 bits representative of 128 bits of each 128 channels ata time. In the polyphase filter of FIGS. 36-37, 32 1-bit input samples(e.g., from the 1-bit ADC output stream at 39.32 MHz) are loaded intothe 32 polyphase filters starting from the bottom and working up. After32 samples are loaded, the polyphase filters run to generate a singleoutput point. Here, 1-bit samples from each of the 128 channels areloaded into the 32 polyphase filters. Thus, the filter structure of FIG.39 runs at 128 times the digital decimation filtering circuit 248 dataoutput rate.

Summation tree 292 adds the results of polyphase filters (LUTs 290)(e.g., 16 bits from E0 is added to 16 bits from E1 to make a 17-bitvalue, and so on) to get the final result of 128 18-bit outputs at 1.23MHz multiplexed on a 157.29 MHz bus.

FIG. 40 is a schematic block diagram of an example of a shift registermemory 286. For example, shift register memory 286 is a two port5-bit×128-bit device (e.g., a static access random memory (SRAM) device)that is written a column (128 bits) at a time and is read a row (5 bits)at a time. Port A is a write only port that has inputs for data A (128bits) and address A (3 bits) that addresses rows. Port B is a read onlyport that has an output for data B (5 bits) and an input for address B(7 bits) that addresses columns. Shift register memory 286 has a latchedoutput.

Each row of shift register memory 286 is the shift register for one ofthe 128 1-bit ADC output channels. Of the 5 taps in each row, 4 taps areactive data read out into the filter structure and the fifth tap isreserved for input data for the next output sample. When the data forthe filter structure of FIG. 39 arrives on the input bus, all 128channels are sampled on that cycle.

Memory is written to each shift memory registry 286 at the samefrequency as the output sample rate (e.g., 1.23 Mz) however, memory isread out into the filter structure at a rate of 128 times the outputsample rate (e.g., 128×1.23 Mz=157.29 MHz). This allows all the 128channels to be processed by the same filter structure, and all theoutput data to be multiplexed onto the same output bus.

FIG. 41 is a schematic block diagram of another embodiment of thedigital decimation filtering circuit 248. FIG. 41 shows a detailedexample of the digital decimation filtering circuit 248 of FIG. 39. Theinput signals of each 128 channels are broken up in order to decimateeach channel by the decimation rate of 32 (e.g., as discussed withreference to FIG. 38). For example, the shift register memory 1 286writes in the first bit (x[0]) from each channel, then writes in the32^(nd) bit (x[32]) from each channel, then writes in the 64^(th) bit(x[64]) from each channel, and then writes in the 96^(th) bit (x[96])from each channel. Shift register memory 1 286 reads out a row of datacontaining x[0], x[32], x[64], and x[96] from each channel at a rate of128 times the write rate (1.23 MHz) in order to filter 128 channelsthrough one filter structure.

The four bits of data from each channel from shift register memory 1 286are filtered through filter E0 (look-up table (LUT) 290) to produce 12816-bit outputs. For example, E₀ filter includes taps h[0], h[32]z⁻¹,h[64]z⁻² and h[96]z⁻³. An input of x[0], x[32], x[64], and x[96]produces a 16-bit filter output y[0] representative of these inputs.Shift register memories 2 through 128 operate similarly to shiftregister memory 1 286. Each 16-bit output from each filter E0-E31 goesthrough summation tree 292 in order to produce one 18-output. Therefore,at the output of digital decimation filtering circuit 128, 128 18-bitvalues are output at a rate of 1.23 MHz multiplexed on a 157.29 MHz bus(e.g., 128×1.23 MHz).

FIG. 42 is an example of a frequency band having n frequency channels282. FIG. 42 is similar to the example of FIG. 35 except now thefrequency band of interest 280 is shown in comparison to the decimationfrequency fd=1.23 MHz after going through the digital decimationfiltering circuit 248 (e.g., the digital decimation filtering circuit248 cut off noise at higher frequencies than 1.23 MHz and reduced thesampling rate to 1.23 MHz). The frequency band of interest 280 begins atf1 and ends at fn. Frequency band of interest 280 includes channels 282f1-fn spaced out at a desired channel spacing 255 (e.g., the data outputrate of 300 Hz or another frequency). With a channel spacing of 300 Hz(e.g., the data output rate), the frequency band of interest is 128×300Hz=38.4 KHz wide (i.e., n×channel spacing 300 Hz). If f1 is at 100 KHz,the frequency band of interest 280 spans from 100 Khz to 138.4 KHz.

FIG. 43 is a schematic block diagram of an embodiment of digitalbandpass filter (BPF) circuit 250. Digital bandpass filter (BPF) circuit250 includes one or more finite impulse response (FIR) filters, one ormore cascaded integrated comb (CIC) filters, one or more infiniteimpulse response (FIR) filters, one or more decimation stages, one ormore fast Fourier transform (FFT) filters, one or more discrete Fouriertransform (DFT) filters, and/or one or more polyphase filters. BPF 250includes a plurality of taps having coefficients set to produce abandpass region approximately centered at the oscillation frequency ofthe analog reference signal (e.g., 100 KHz) and having a bandwidth tunedfor filtering a pure tone (e.g., f1). BPF 250 has a frequency responseH(z).

Digital BPF circuit 250 takes the output of the digital decimationfiltering circuit 248 (e.g., the 18-bit output at approximately 1.23 MHzrepresentative of signal s1 at frequency f1) and shifts to the bandpassfor frequency f1 (e.g., 100 KHz). When the output of the digitaldecimation filtering circuit 248 includes n 18-bit outputs fromdifferent channels (e.g., the analog input signal includes pure tonecomponents f1-fn of FIG. 42), a digital BPF circuit is needed for eachoutput to isolate each pure tone component.

Digital BPF circuit 250 applies a very narrow bandpass filter andoutputs an affect value 254 (si) having real and imaginary components atthe output frequency of 300 Hz. Because embedded data is a sinusoid(e.g., a pure tone) the desired information is at frequency f1 and basedon magnitude and/or phase. Therefore, the bandpass filter can be verynarrow (e.g., less than 0.05 the channel spacing (e.g., 10 Hz)) tocapture the desired signal.

FIG. 44 is an example frequency response H(z) of digital bandpass filter(BPF) circuit 250. As an example, the digital BPF circuit 250 is adiscrete Fourier transform (DFT) filter with length N. For example,digital BPF circuit 250 has a length 4096 in order to filter 4096 18-bitinputs to produce 1 48-output. A sin x/x (e.g., or “sinc”) frequencyresponse is shown. The sinc frequency response has a “notch” response(e.g., it can reject the line frequency when set to that frequency).With a sampling frequency of 1.23 MHz and length 4096, the frequency bin(i.e., intervals between samples in frequency domain) resolution is 1.23MHz/4096=300 Hz. The notch position is also directly related to theoutput data rate. As shown, the sinc frequency response is equal to zeroat integer multiples of the output data rate of 300 Hz (e.g., 600 Hz,900 Hz, 1200 Hz, and so on).

FIG. 45 is an example frequency response H(z) of a digital bandpassfilter (BPF) circuit 250. A finite impulse response (FIR) filter has asin x/x (e.g., or “sinc”) frequency response as shown. The sincfrequency response has a “notch” response (e.g., it can reject the linefrequency when set to that frequency). With the signal s1 shifted tobandpass, a very narrow bandpass filter can be applied. For example, abandpass filter of 10 Hz with center frequency 5 Hz is applied toisolate the pure tone. As shown, the first notch position in FIG. 45 islocated at 10 Hz with a center frequency of 5 Hz.

FIGS. 46A-46D are examples of processing a signal by digital bandpassfilter (BPF) circuit 1 250. BPF circuit 1 250 includes a plurality oftaps having coefficients set to produce a bandpass region approximatelycentered at the oscillation frequency of an analog reference signal fors1 (e.g., 100 KHz) and having a bandwidth tuned for filtering a digitalsignal having frequency components at f1, f2, and f3. In FIG. 46A,digital BPF circuit 1 250 receives the output of the digital decimationfiltering circuit 248 (e.g., the 18-bit output at approximately 1.23 MHzrepresentative of signals s1 at frequency f1, s2 at frequency f2, and s3at frequency f3).

In FIG. 46B digital BPF circuit 1 250 shifts the 18-bit output atapproximately 1.23 MHz representative of signals s1 at frequency f1, s2at frequency f2, and s3 at frequency f3 to the bandpass for frequency f1(e.g., 100 KHz). For example, s1 is now at 0 Hz and s2 and s3 are spacedout evenly from s1 (e.g., at 300 Hz and 600 Hz).

In FIG. 46C, digital BPF circuit 1 250 applies a very narrow bandpassfilter to isolate s1. Because the embedded data is a sinusoid (e.g., apure tone) the desired information is at frequency f1 (e.g., 0 Hz) andbased on magnitude and/or phase. Therefore, the bandpass filter can bevery narrow (e.g., less than 0.05 the channel spacing (e.g., 10 Hz)) tocapture the desired signal.

In FIG. 46D, digital BPF circuit 1 250 outputs an affect value 254 (s1)having real and imaginary components at the output frequency of 300 Hz.The affect value 254 (s1) is 48 bits with a 24-bit real part and a24-bit imaginary part.

FIGS. 47A-47D are examples of processing a signal by digital bandpassfilter (BPF) circuit 2 250. BPF circuit 2 250 includes a plurality oftaps having coefficients set to produce a bandpass region approximatelycentered at the oscillation frequency of an analog reference signal fors2 (e.g., 100.3 KHz) and having a bandwidth tuned for filtering a puretone (e.g., f2). In FIG. 47A, digital BPF circuit 2 250 receives theoutput of the digital decimation filtering circuit 248 (e.g., the 18-bitoutput at approximately 1.23 MHz representative of signals s1 atfrequency f1, s2 at frequency f2, and s3 at frequency f3).

In FIG. 47B digital BPF circuit 2 250 shifts the 18-bit output atapproximately 1.23 MHz representative of signals s1 at frequency f1, s2at frequency f2, and s3 at frequency f3 to the bandpass for frequency f2(e.g., 100.3 KHz). For example, s2 is now at 0 Hz and s3 at 300 Hz. S1may fold over and be aligned with s3 or another frequency.

In FIG. 47C, digital BPF circuit 2 250 applies a very narrow bandpassfilter to isolate s2. Because the embedded data is a sinusoid (e.g., apure tone) the desired information is at frequency f2 (e.g., 0 Hz) andbased on magnitude and/or phase. Therefore, the bandpass filter can bevery narrow (e.g., less than 0.05 the channel spacing (e.g., 10 Hz)) tocapture the desired signal.

In FIG. 47D, digital BPF circuit 2 250 outputs an affect value 254 (s2)having real and imaginary components at the output frequency of 300 Hz.The affect value 254 (s2) is 48 bits with a 24-bit real part and a24-bit imaginary part.

FIG. 48 is a schematic block diagram of an embodiment of digitalbandpass filter (BPF) circuit 250. Digital BPF circuit 250 includes oneor more finite impulse response (FIR) filters, one or more cascadedintegrated comb (CIC) filters, one or more infinite impulse response(FIR) filters, one or more decimation stages, one or more fast Fouriertransform (FFT) filters, one or more discrete Fourier transform (DFT)filters, and/or one or more polyphase filters.

Using a DFT filter example, BPF circuit 250 receives n channels ofoutput from the digital decimation filtering circuit 248 (e.g., 12818-bit outputs at approximately 1.23 MHz on a 16-bit bus running atapproximately 157.29 MHz) and computes discrete Fourier transforms witha length N (e.g., to have a frequency response that synchronizesoutputs) on every channel at two frequencies at a time (e.g., aself-capacitance frequency and a pen frequency). Digital BPF circuit 250runs at twice the speed of the input bus (e.g., 2²⁰×300 Hz=314.57 MHz)to compute two frequencies for each input.

Discrete Fourier transforms (DFTs) transform a sequence of complexnumbers into another sequence of complex numbers. Each 18-bit inputreceived represents a complex number at a frequency. An 18-bit input isfed to two separate multipliers where the input is multiplied by eitherthe real or imaginary coefficients of the DFT. The real and imaginaryparts of a DFT are pre-computed for each frequency by a coefficientprocessor. The coefficient processor will be discussed in more detailwith reference to FIG. 52.

For each real and imaginary part of the input, BPF circuit 250 applies amultiply accumulate function. For example, real part coefficients 294are multiplied by the input and accumulated by accumulators 302. Whenthe final result is computed for all outputs (e.g., 128 channels×2frequencies=256 outputs) the final result is shifted to output buffer304 and real component value 298 (e.g., 24 bits) is output at 300 Hz.Likewise, imaginary part coefficients 296 are multiplied by the inputand accumulated by accumulators 302. When the final result is computedfor all outputs (128 channels×2 frequencies=256 outputs) the finalresult is shifted to output buffer 304 and imaginary component value 300(e.g., 24 bits) is output at 300 Hz. Therefore, the BPF circuit filtersN outputs from digital decimation filtering circuit 248 to produce 1output with a real component value 298 and an imaginary component value300 and operates at twice the speed of the input in order to output aresult for two frequencies per channel.

FIG. 49 is a schematic block diagram of another embodiment of analog todigital conversion circuit 246 that includes analog to digital converter(ADC) circuits 1-n 258, digital filtering decimation circuit 248, 1^(st)bandpass filter (BPF) circuit 250, 2^(nd) BPF circuit 250, coefficientprocessor 306, and processing module 252.

ADC circuits 1-n 258 may be ADC converter 212 of drive sense circuit 28of previous Figures and/or any conventional ADC (e.g., a flash ADC, asuccessive approximation ADC, a ramp-compare ADC, a Wilkinson ADC, anintegrating ADC, and/or a delta encoded ADC). ADC circuit 258 may beimplemented by combination of a 1-bit ADC sigma-delta modulator and thedigital decimation filtering circuit 248 (e.g., a sigma delta ADC). FIG.49 operates in accordance with previous examples except that 1^(st) BPFcircuit 250 is operable to process outputs from digital filteringdecimation circuit 248 where frequencies are known and 2nd BPF circuit250 is operable to process outputs from digital filtering decimationcircuit 248 where frequencies are selectable by the processing module252 according to regions of interest (ROI).

For example, mutual capacitance is measured at two points per n channels(e.g., 256 frequencies where n is equal to 128). Processing module 252chooses which cross points are sampled based on given crossing points orat specific points based on the results of the of self-frequencies andpen frequencies which are always on. Processing module 252 inputs theselected frequencies 310 into 2^(nd) BPF 250 and coefficient processer306 pre-computes coefficients for 2^(nd) BPF 250 based on the selectedfrequencies. 2^(nd) BPF 250 will be discussed in more detail withreference to FIG. 51.

Self-capacitance is measured at one point (e.g., one frequency) per nelectrodes of a touch screen (e.g., 128 electrodes, where n is equal to128) and pen capacitance is measured at one point (e.g., one frequencyon the same n electrodes). These frequencies are known to the systemtherefore processing module 252 inputs the known frequencies 308 intocoefficient processer 306 to pre-compute coefficients for 1st BPF 250.1^(st) BPF 250 will be discussed in more detail with reference to FIG.50.

1^(st) BPF 250 outputs 1^(st) affect values 254 representative of selfand pen capacitance values (e.g., 1^(st) BPF 250 runs at twice the speedof the input bus to filter two frequencies per channel as discussed inFIG. 48). 2^(nd) BPF 250 outputs 2^(nd) affect values 254 representativeof mutual capacitance values. Processing module 252 interprets theimaginary and real components of the affect values 254 to produce dataoutput 256. Affect values 254 are vectors (i.e., phasor complex numbers)having a real component and an imaginary component representing asinusoidal function that has a peak magnitude (i.e., amplitude) anddirection (i.e., phase). For example, affect values 254 are 48-bitvalues each having a 24-bit real component and a 24-bit imaginarycomponent. In the complex domain, voltages and currents are phasors andall resistances, capacitances, and inductances are replaced with compleximpedances (e.g., Z_(R)=R, Z_(L)=jfL, and Z_(C)=1/(jfC)=−j/(fC)).

Since the impedance of a channel is primarily based on its capacitance(self, pen, and/or mutual), as the frequency increases for a fixedcapacitance, the impedance decreases based on ½πfC, where f is thefrequency and C is the capacitance. Since voltage (V)=current(I)*impedance (Z), the processing module 252 determines a capacitance orother impedance value from voltage and current vectors of the affectvalue 254 (e.g., a decrease in impedance increases the voltage for aconstant current, increases the current for a constant voltage, orincreases both voltage and current of the signal component). Theincreasing and/or decreasing of impedance is representative of the inputdata. The impedance value or change in impedance value determined isoutput as data output 256 at the example output rate of 300 Hz.

FIG. 50 is a schematic block diagram of an embodiment of a firstbandpass filter (BPF) circuit 250. 1^(st) BPF circuit 250 is one or morediscrete Fourier transform (DFT) or fast Fourier transform (FFT) filtersthat receives n (e.g., 128) channels of output from the digitaldecimation filtering circuit 248 (e.g., 128 18-bit outputs atapproximately 1.23 MHz on a 16-bit bus running at approximately 157.29MHz) and computes discrete Fourier transforms with a length N=4096 onevery channel at two frequencies at a time (e.g., a self-capacitancefrequency and a pen frequency). Digital BPF circuit 250 runs at twicethe speed of the input bus (e.g., 2²⁰×300 Hz=314.57 MHz) to compute twofrequencies for each input.

Discrete Fourier transforms transform a sequence of N complex numbers(e.g., {x_(n)}=x₀, x₁, . . . , x_(N-1)) into another sequence of complexnumbers (e.g., {X_(k)}=X₀, X₁, . . . , X_(N-1)) where

$X_{k} = {\sum\limits_{n = 0}^{N - 1}{x_{n}w_{n}e^{\frac{{- i}\; 2\pi}{N}kn}}}$which can also be expressed as

$X_{k} = {\sum\limits_{n = 0}^{N - 1}{x_{n}{w_{n}\left\lbrack {{\cos\left( \frac{2\pi kn}{N} \right)} - {i \cdot {\sin\left( \frac{2\pi kn}{N} \right)}}} \right\rbrack}}}$according to Euler's formula.

For each 18-bit input, the real and imaginary parts of

$w_{n}e^{\frac{{- i}\; 2\pi}{N}kn}$are pre-computed for each frequency by coefficient processor 306.Coefficient processor 306 will be discussed in more detail withreference to FIG. 52.

For example, the real part coefficients w_(n) cos(2πω_(s)n) (for aself-frequency (fs) input ω_(s), where

$\left. {\frac{k}{N} = \omega_{s}} \right)$and w_(n) cos(2πω_(p)n) (for a pen-frequency (fp) input ω_(p), where

$\left. {\frac{k}{N} = \omega_{p}} \right)$are precomputed by coefficient processor 306 and multiplexed to bemultiplied with the correct 18-bit output from digital decimationfiltering circuit 248 shown on the left side. Imaginary partcoefficients −w_(n) sin(2πω_(s)n) (for a self-frequency (fs) inputω_(s), where

$\left. {\frac{k}{N} = \omega_{s}} \right)$and −w_(n) sin(2πω_(p)n) (for a pen-frequency (fp) input ω_(p), where

$\left. {\frac{k}{N} = \omega_{p}} \right)$are precomputed by coefficient processor 306 and multiplexed to bemultiplied with the correct 18-bit output from digital decimationfiltering circuit 248 shown on the right side.

1^(st) BPF circuit 250 includes 4 18×18 multipliers (e.g., 2 per side,per frequency) for multiplying 18-bits from the digital decimationfiltering circuit 248 with 18-bits from coefficient processor. 1^(st)BPF circuit 250 includes 4 30-bit signed accumulators 302 (e.g., 2 perside, per frequency) with 256 output registers for adding the multipliedvalues. 1^(st) BPF circuit 250 further includes 4 256×24-bit outputbuffers 304 (e.g., 2 per side, per frequency). For example, outputbuffers 304 are two port static access random memory (SRAM) with a writeonly port and a read only port.

1^(st) BPF circuit 250 computes the final product for all 256 18-bitvalues received (e.g., 128 channels×2 frequencies) and shifts the finalresults to output buffers 304. Output buffers 304 output a 24-bit realcomponent values 298 and a 24-bit imaginary component values 300 outputat 300 Hz to the processing module 252.

FIG. 51 is a schematic block diagram of an embodiment of a secondbandpass filter (BPF) circuit 250. 2^(nd) BPF circuit 250 is one or morediscrete Fourier transform (DFT) or fast Fourier transform (FFT) filtersthat receives n (e.g., 128) channels of output from the digitaldecimation filtering circuit 248 (e.g., 128 18-bit outputs atapproximately 1.23 MHz on a 16-bit bus running at approximately 157.29MHz) and computes discrete Fourier transforms with a length N=4096 onevery channel at two frequencies at a time (e.g., two mutual frequenciesper channel). Digital BPF circuit 250 runs at twice the speed of theinput bus (e.g., 2²⁰×300 Hz=314.57 MHz) to compute two frequencies foreach input.

2nd BPF circuit 250 is operable to process outputs from digitalfiltering decimation circuit 248 where frequencies are selectableaccording to regions of interest (ROI). For example, mutual capacitanceis measured at two points per n channels (e.g., 256 frequencies where nis equal to 128). Processing module 252 chooses which cross points aresampled based on given crossing points or at specific points based onthe results of the of self-frequencies and pen frequencies which arealways on. Processing module 252 inputs the selected frequencies intooutput map 316 of 2^(nd) BPF 250.

For example, output map 316 is a 256×12-bit two port a static accessrandom memory (SRAM) which has a write only port interfaced with theprocessing module and a read only port. The processing module fills outthe 256-entry output map 316 with a 12-bit address for each frequency(e.g., 7-bit address 326 for channel number and 5-bit address 328 forthe frequency index). Input buffer 312 is double buffered (in comparisonto the first BPF circuit 250) so the channels can load sequentially andclock out randomly. For example, input buffer 312 is a 256×18-bit twoport SRAM with a write only port and a read only port. Because the inputis double buffered, 2nd BPF circuit 250 will have one more cycle oflatency (e.g., at 300 Hz) than the 1^(st) BPF circuit 250.

Counter 314 clocks through the 256 cross points at each summation step.Each cross point selects a line and coefficients corresponding to one ofthe frequencies computed by the coefficient processor 306. Coefficientprocesser 306 pre-computes coefficients for 2^(nd) BPF 250 based on theselected frequencies. For example, there are 34 possible frequencyvalues for mutual frequency. Therefore, coefficient processor 306computes real and imaginary coefficients for each of the 34possibilities and stores the values in look-up tables.

Coefficient processer 306 inputs real coefficients into coefficientlookup table (LUT) 318 and imaginary coefficients into coefficient LUT320. Each coefficient LUT 318 and 320 is a 34×18-bit two port SRAM witha read only port and a write only port. The 5-bit address 328corresponding to a selected frequency is input to each coefficient LUT318 and 320 and is used to select the correct coefficients for theinput. Coefficient LUT 318 inputs an 18-bit real coefficient to themultiplier to be multiplied by the 18-bit input value from digitalfiltering decimation circuit 248. Coefficient LUT 320 inputs an 18-bitimaginary coefficient to the second multiplier on the right of theschematic to be multiplied by the 18-bit input value from digitalfiltering decimation circuit 248.

The rest of 2^(nd) BPF 250 operates similarly to 1^(st) BPF circuit 250.2nd BPF circuit 250 includes 4 18×18 multipliers (e.g., 2 per side, perfrequency) for multiplying 18-bits from the digital decimation filteringcircuit 248 with 18-bits from coefficient LUTs 318 and 320. 2^(nd) BPFcircuit 250 includes 4 30-bit signed accumulators 302 (e.g., 2 per side,per frequency) with 256 output registers for adding the multipliedvalues. 2^(nd) BPF circuit 250 further includes 4 256×24-bit outputbuffers 304 (e.g., 2 per side, per frequency). For example, outputbuffers 304 are two port SRAM with a write only port and a read onlyport. 2^(nd) BPF circuit 250 computes the final product for all 25618-bit values received (e.g., 128 channels×2 frequencies) and shifts thefinal results to output buffers 304. Output buffers 304 output the24-bit real component values 322 and the 24-bit imaginary componentvalues 324 output at 300 Hz to the processing module.

FIG. 52 is a schematic block diagram of an embodiment of a coefficientprocessor 306. Coefficient processor 306 includes a “nk” latch, a +1024,a function multiplexer, cosine lookup table 330, frequency lookup table332, a 0.5-0.5×, a “w_(n)” latch, counter 334, two multiplexers, and acoefficient processor multiplier 336. Frequency lookup table 332 is a34×12-bit two port a static access random memory (SRAM) that is filledwith the 34 possible frequency options by the processing module. Cosinelookup table 330 is a 1.17 fixed point 4096×16-bit lookup table (e.g.,read only memory (ROM)). Coefficient processor multiplier 336 is an18×18-bit signed multiplier.

As discussed previously, discrete Fourier transforms (DFTs) transform asequence of N complex numbers (e.g., {x_(n)}=x₀, x₁, . . . , x_(N-1))into another sequence of complex numbers (e.g., {X_(k)}=X₀, X₁, . . . ,X_(N-1)) where

$X_{k} = {\sum\limits_{n = 0}^{N - 1}{x_{n}w_{n}e^{\frac{{- i}\; 2\pi}{N}kn}}}$which can also be expresses as

$X_{k} = {\sum\limits_{n = 0}^{N - 1}{x_{n}{w_{n}\left\lbrack {{\cos\left( \frac{2\pi kn}{N} \right)} - {i \cdot {\sin\left( \frac{2\pi kn}{N} \right)}}} \right\rbrack}}}$according to Euler's formula.

For each 18-bit input, the real and imaginary parts of

$w_{n}e^{\frac{{- i}\; 2\pi}{N}kn}$are pre-computed for each frequency by coefficient processor 306.

For example, there are 34 possible frequency options that every inputinto the BPF circuits 250 could be. Each real part coefficient w_(n)cos(2πωn) (for a frequency input ω, where

$\left. {\frac{k}{N} = \omega} \right)$and each imaginary part coefficient −w_(n) sin(2πωn) (for a frequencyinput ω, where

$\left. {\frac{k}{N} = \omega} \right)$for each of the 34 frequency options is precomputed by coefficientprocessor 306.

Coefficient processor 306 runs at a rate of n that is one greater thanthe BPF circuits 250 so that it is always one cycle ahead of the BPFscircuits 250. For each value of n, (which runs at the input sample rateof 1.23 MHz) the coefficient processor 306 makes the followingcomputations: 1) w_(n)=0.5-0.5 cos(2πn/N) and for each of the 34frequency possibilities (k): 1) kn mod 4096, 2) w_(n) cos(2πkn/N), and3)−w_(n) sin(2πkn/N). To complete this, coefficient processor multiplier336 runs at n times the input sample rate (e.g., 128×1.23 MHz=157.29 MHzwhen n is 128).

Counter 334 goes from 0 to 4095 and inputs the value of “n” into thefunction multiplexer and the multiplexer connected with w_(n) latch. Thekn latch inputs the kn value to the function multiplexer for thecos(2πkn/N) and −sin(2πkn/N) functions. A kn plus 1024 value is alsoinput to the function multiplexer. Based on the clock cycle, a functionis selected (e.g., cos(2πkn/N), −sin(2πkn/N), or cos(2πn/N)). Theselected function is entered into the cosine lookup table 330 to lookupthe particular value for that function.

If the function selected does not include k, the particular value fromcosine lookup table 330 is fed to the 0.5-0.5× where the value forw_(n)=0.5-0.5 cos(2πn/N) is computed and outputted to the w_(n) latch.The w_(n) latch outputs a value and is multiplexed with counter 334outputs. When the function does include k, the selected function isentered into the cosine lookup table 330 to lookup the particular valuefor that function. The particular value is then input into themultiplexer with an output (k) from the frequency lookup table 332. Onone clock cycle, the particular value for either for cos(2πkn/N) or−sin(2πkn/N) is input to the coefficient processor multiplier 336 to bemultiplied by the w_(n) value. The computed coefficients are convertedto 18 bits then output to the BPF circuits 250.

On a different clock cycle, a value of k from the frequency lookup table332 is input to the coefficient processor multiplier 336 to bemultiplied with the value n from counter 334. The kn value is input intothe kn latch for the next set of computations.

FIG. 53 is a schematic block diagram of another embodiment of analog todigital conversion circuit 246 that includes analog to digital converter(ADC) circuits 1-n 258, digital filtering decimation circuit 248, penbandpass filter (BPF) circuit 338, self BPF circuit 340, coefficientprocessor 306, mutual BPF circuit 342, and processing module 252.

FIG. 53 operates similarly to the example of FIG. 49 except two BPFfilters running at the speed of the input bus (e.g., pen BPF circuit 338and self BPF circuit 340) replace one BPF circuit 250 (e.g., 1^(st) BPFcircuit 250) running at twice the speed of the input bus in order tocompute two frequencies at a time. Pen BPF circuit 338 and self BPFcircuit 340 process outputs from digital filtering decimation circuit248 where frequencies are known (e.g., self and pen measurements arealways on) and mutual BPF circuit 342 is operable to process outputsfrom digital filtering decimation circuit 248 where frequencies areselectable according to regions of interest (ROI).

For example, mutual capacitance is measured at two points per nelectrodes (e.g., 256 frequencies where n is equal to 128). Processingmodule 252 chooses which cross points are sampled based on givencrossing points or at specific points based on the results of theself-frequencies and pen frequencies which are always on. Processingmodule 252 inputs a selection of frequencies 310 into mutual BPF circuit342 and coefficient processer 306 pre-computes coefficients for mutualBPF circuit 342 based on the selected frequencies.

Self-capacitance is measured at one point (e.g., frequency) per nelectrodes and pen capacitance is measured at one point per (e.g., onefrequency) n electrodes. These frequencies are known to the systemtherefore processing module 252 inputs the known frequencies 308 intocoefficient processer 306 to pre-compute coefficients for pen BPFcircuit 338 and self BPF circuit 340.

Pen BPF circuit 338 outputs pen affect values representative of pencapacitance values. Self BPF circuit 340 outputs self affect valuesrepresentative of self capacitance values. Mutual BPF circuit 342outputs mutual affect values representative of mutual capacitancevalues. Processing module 252 interprets the imaginary and realcomponents of the affect values to produce data output 256 at 300 Hz.Affect values 254 are vectors (i.e., phasor complex numbers) having areal component and an imaginary component representing a sinusoidalfunction that has a peak magnitude (i.e., amplitude) and direction(i.e., phase). For example, affect values 254 are 48-bit values eachhaving a 24-bit real component and a 24-bit imaginary component. In thecomplex domain, voltages and currents are phasors and all resistances,capacitances, and inductances are replaced with complex impedances(e.g., Z_(R)=R, Z_(L)=jfL, and Z_(C)=1/(jfC)=−j/(fC)).

Since the impedance of a channel is primarily based on its capacitance(self, pen, and/or mutual), as the frequency increases for a fixedcapacitance, the impedance decreases based on ½πfC, where f is thefrequency and C is the capacitance. Since voltage (V)=current(I)*impedance (Z), the processing module 252 determines a capacitance orother impedance value from voltage and current vectors of the affectvalue (e.g., a decrease in impedance increases the voltage for aconstant current, increases the current for a constant voltage, orincreases both voltage and current of the signal component). Theincreasing and/or decreasing of impedance is representative of the inputdata. The impedance value or change in impedance value determined isoutput as data output 256 at the example output rate of 300 Hz.

FIG. 54 is a schematic block diagram of an embodiment of processingmodule 252 controls within the analog to digital conversion circuit 246.Analog to digital conversion circuit 246 is a confined datacommunication system in all variables are set by the processing module252 and controlled for desired data processing. Processing module 252 isoperable to control every stage of analog to digital conversion circuit246 in order to produce the desired output 256.

For example, processing module 252 sets the frequency and waveform foreach oscillating reference signal via reference generation circuit 344(e.g., reference signal generator 149) to produce analog referencesignals 346. DC component input data 348 is embedded in each analogreference signal 346. Processing module 252 also sets the sampling rateof ADC 258. ADC 258 processes the analog signal containing the analogreference signal and the DC component and outputs representative signal350 to the digital filtering stages 352 (e.g., digital decimationfiltering circuit 248 and digital BPF circuit 250).

Processing module 252 determines the stages (e.g., taps) of each filter,the sampling frequencies, the filter bandwidth, and any other desiredfilter parameters. Processing module 252 determines digital filteringparameters based on a desired output rate, desired linearity, and otherfactors. Processing module 252 inputs known frequencies and mutualfrequency selections into the coefficient processor for digital BPFfilters.

The digital filtering stage 352 produces an affect value 254 to beinterpreted by the processing module 252 at the data processing 354stage. Processing module 252 sets data interpretation parameters basedon the data output rate and the nature of the input data 348.

For example, input data 348 may be communicating one or more of current(I), voltage (V), or impedance (Z) changes. For example, if the input isa voltage measurement with a constant current, processing module 252 cananalyze the voltage change to determine an impedance change value. Basedon the data interpretation parameters, processing module 252 interpretsaffect value 254 and produces processed output data 256.

As may be used herein, the terms “substantially” and “approximately”provide an industry-accepted tolerance for its corresponding term and/orrelativity between items. For some industries, an industry-acceptedtolerance is less than one percent and, for other industries, theindustry-accepted tolerance is 10 percent or more. Other examples ofindustry-accepted tolerance range from less than one percent to fiftypercent. Industry-accepted tolerances correspond to, but are not limitedto, component values, integrated circuit process variations, temperaturevariations, rise and fall times, thermal noise, dimensions, signalingerrors, dropped packets, temperatures, pressures, material compositions,and/or performance metrics. Within an industry, tolerance variances ofaccepted tolerances may be more or less than a percentage level (e.g.,dimension tolerance of less than +/−1%). Some relativity between itemsmay range from a difference of less than a percentage level to a fewpercent. Other relativity between items may range from a difference of afew percent to magnitude of differences.

As may also be used herein, the term(s) “configured to”, “operablycoupled to”, “coupled to”, and/or “coupling” includes direct couplingbetween items and/or indirect coupling between items via an interveningitem (e.g., an item includes, but is not limited to, a component, anelement, a circuit, and/or a module) where, for an example of indirectcoupling, the intervening item does not modify the information of asignal but may adjust its current level, voltage level, and/or powerlevel. As may further be used herein, inferred coupling (i.e., where oneelement is coupled to another element by inference) includes direct andindirect coupling between two items in the same manner as “coupled to”.

As may even further be used herein, the term “configured to”, “operableto”, “coupled to”, or “operably coupled to” indicates that an itemincludes one or more of power connections, input(s), output(s), etc., toperform, when activated, one or more its corresponding functions and mayfurther include inferred coupling to one or more other items. As maystill further be used herein, the term “associated with”, includesdirect and/or indirect coupling of separate items and/or one item beingembedded within another item.

As may be used herein, the term “compares favorably”, indicates that acomparison between two or more items, signals, etc., provides a desiredrelationship. For example, when the desired relationship is that signal1 has a greater magnitude than signal 2, a favorable comparison may beachieved when the magnitude of signal 1 is greater than that of signal 2or when the magnitude of signal 2 is less than that of signal 1. As maybe used herein, the term “compares unfavorably”, indicates that acomparison between two or more items, signals, etc., fails to providethe desired relationship.

As may be used herein, one or more claims may include, in a specificform of this generic form, the phrase “at least one of a, b, and c” orof this generic form “at least one of a, b, or c”, with more or lesselements than “a”, “b”, and “c”. In either phrasing, the phrases are tobe interpreted identically. In particular, “at least one of a, b, and c”is equivalent to “at least one of a, b, or c” and shall mean a, b,and/or c. As an example, it means: “a” only, “b” only, “c” only, “a” and“b”, “a” and “c”, “b” and “c”, and/or “a”, “b”, and “c”.

As may also be used herein, the terms “processing module”, “processingcircuit”, “processor”, “processing circuitry”, and/or “processing unit”may be a single processing device or a plurality of processing devices.Such a processing device may be a microprocessor, micro-controller,digital signal processor, microcomputer, central processing unit, fieldprogrammable gate array, programmable logic device, state machine, logiccircuitry, analog circuitry, digital circuitry, and/or any device thatmanipulates signals (analog and/or digital) based on hard coding of thecircuitry and/or operational instructions. The processing module,module, processing circuit, processing circuitry, and/or processing unitmay be, or further include, memory and/or an integrated memory element,which may be a single memory device, a plurality of memory devices,and/or embedded circuitry of another processing module, module,processing circuit, processing circuitry, and/or processing unit. Such amemory device may be a read-only memory, random access memory, volatilememory, non-volatile memory, static memory, dynamic memory, flashmemory, cache memory, and/or any device that stores digital information.Note that if the processing module, module, processing circuit,processing circuitry, and/or processing unit includes more than oneprocessing device, the processing devices may be centrally located(e.g., directly coupled together via a wired and/or wireless busstructure) or may be distributedly located (e.g., cloud computing viaindirect coupling via a local area network and/or a wide area network).Further note that if the processing module, module, processing circuit,processing circuitry and/or processing unit implements one or more ofits functions via a state machine, analog circuitry, digital circuitry,and/or logic circuitry, the memory and/or memory element storing thecorresponding operational instructions may be embedded within, orexternal to, the circuitry comprising the state machine, analogcircuitry, digital circuitry, and/or logic circuitry. Still further notethat, the memory element may store, and the processing module, module,processing circuit, processing circuitry and/or processing unitexecutes, hard coded and/or operational instructions corresponding to atleast some of the steps and/or functions illustrated in one or more ofthe Figures. Such a memory device or memory element can be included inan article of manufacture.

One or more embodiments have been described above with the aid of methodsteps illustrating the performance of specified functions andrelationships thereof. The boundaries and sequence of these functionalbuilding blocks and method steps have been arbitrarily defined hereinfor convenience of description. Alternate boundaries and sequences canbe defined so long as the specified functions and relationships areappropriately performed. Any such alternate boundaries or sequences arethus within the scope and spirit of the claims. Further, the boundariesof these functional building blocks have been arbitrarily defined forconvenience of description. Alternate boundaries could be defined aslong as the certain significant functions are appropriately performed.Similarly, flow diagram blocks may also have been arbitrarily definedherein to illustrate certain significant functionality.

To the extent used, the flow diagram block boundaries and sequence couldhave been defined otherwise and still perform the certain significantfunctionality. Such alternate definitions of both functional buildingblocks and flow diagram blocks and sequences are thus within the scopeand spirit of the claims. One of average skill in the art will alsorecognize that the functional building blocks, and other illustrativeblocks, modules and components herein, can be implemented as illustratedor by discrete components, application specific integrated circuits,processors executing appropriate software and the like or anycombination thereof.

In addition, a flow diagram may include a “start” and/or “continue”indication. The “start” and “continue” indications reflect that thesteps presented can optionally be incorporated in or otherwise used inconjunction with one or more other routines. In addition, a flow diagrammay include an “end” and/or “continue” indication. The “end” and/or“continue” indications reflect that the steps presented can end asdescribed and shown or optionally be incorporated in or otherwise usedin conjunction with one or more other routines. In this context, “start”indicates the beginning of the first step presented and may be precededby other activities not specifically shown. Further, the “continue”indication reflects that the steps presented may be performed multipletimes and/or may be succeeded by other activities not specificallyshown. Further, while a flow diagram indicates a particular ordering ofsteps, other orderings are likewise possible provided that theprinciples of causality are maintained.

The one or more embodiments are used herein to illustrate one or moreaspects, one or more features, one or more concepts, and/or one or moreexamples. A physical embodiment of an apparatus, an article ofmanufacture, a machine, and/or of a process may include one or more ofthe aspects, features, concepts, examples, etc. described with referenceto one or more of the embodiments discussed herein. Further, from figureto figure, the embodiments may incorporate the same or similarly namedfunctions, steps, modules, etc. that may use the same or differentreference numbers and, as such, the functions, steps, modules, etc. maybe the same or similar functions, steps, modules, etc. or differentones.

While the transistors in the above described figure(s) is/are shown asfield effect transistors (FETs), as one of ordinary skill in the artwill appreciate, the transistors may be implemented using any type oftransistor structure including, but not limited to, bipolar, metal oxidesemiconductor field effect transistors (MOSFET), N-well transistors,P-well transistors, enhancement mode, depletion mode, and zero voltagethreshold (VT) transistors.

Unless specifically stated to the contra, signals to, from, and/orbetween elements in a figure of any of the figures presented herein maybe analog or digital, continuous time or discrete time, and single-endedor differential. For instance, if a signal path is shown as asingle-ended path, it also represents a differential signal path.Similarly, if a signal path is shown as a differential path, it alsorepresents a single-ended signal path. While one or more particulararchitectures are described herein, other architectures can likewise beimplemented that use one or more data buses not expressly shown, directconnectivity between elements, and/or indirect coupling between otherelements as recognized by one of average skill in the art.

The term “module” is used in the description of one or more of theembodiments. A module implements one or more functions via a device suchas a processor or other processing device or other hardware that mayinclude or operate in association with a memory that stores operationalinstructions. A module may operate independently and/or in conjunctionwith software and/or firmware. As also used herein, a module may containone or more sub-modules, each of which may be one or more modules.

As may further be used herein, a computer readable memory includes oneor more memory elements. A memory element may be a separate memorydevice, multiple memory devices, or a set of memory locations within amemory device. Such a memory device may be a read-only memory, randomaccess memory, volatile memory, non-volatile memory, static memory,dynamic memory, flash memory, cache memory, and/or any device thatstores digital information. The memory device may be in a form asolid-state memory, a hard drive memory, cloud memory, thumb drive,server memory, computing device memory, and/or other physical medium forstoring digital information.

While particular combinations of various functions and features of theone or more embodiments have been expressly described herein, othercombinations of these features and functions are likewise possible. Thepresent disclosure is not limited by the particular examples disclosedherein and expressly incorporates these other combinations.

What is claimed is:
 1. A method comprises: converting, by n analog todigital converter (ADC) circuits of an analog to digital conversioncircuit, n analog signals into n first digital signals having a firstdata rate frequency, wherein an analog signal of the n analog signalsincludes an oscillation frequency and a set of pure tone components, andwherein n is an integer greater than 1; converting, by n digitaldecimation filtering circuits of the analog to digital conversioncircuit, the n first digital signals into n second digital signalshaving a second data rate frequency; and converting, by n digitalbandpass filter (BPF) circuits of the analog to digital conversioncircuit, the n second digital signals into a plurality of outbounddigital signals having a third data rate frequency, wherein a digitalBPF circuit of the n digital BPF circuits includes a plurality of tapshaving a plurality of coefficients, wherein the plurality ofcoefficients for the plurality of taps is set to produce a bandpassregion approximately centered at the oscillation frequency of the analogsignal and having a bandwidth tuned for filtering a pure tone componentof the set of pure tone components, wherein the first data ratefrequency is a first integer multiple of the third data rate frequency,wherein the second data rate frequency is a second integer multiple ofthe third data rate frequency, and wherein the first data rate frequencyis greater than the second data rate frequency.
 2. The method of claim1, wherein the first data rate frequency is a third integer multiple ofthe second data rate frequency.
 3. The method of claim 1, wherein an ADCcircuit of the n ADC circuits includes a digital decimation filteringcircuit of the n digital decimation filtering circuits.
 4. The method ofclaim 1, wherein a first digital signal of then first digital signals isa 1-bit digital data stream at the first data rate frequency.
 5. Themethod of claim 1 further comprises: converting, by a first digital BPFcircuit of the digital BPF circuit, a second digital signal of the nsecond digital signals into a first outbound digital signal of theplurality of outbound digital signals, wherein the first digital BPFcircuit includes a first plurality of taps having a first plurality ofcoefficients, wherein the first plurality of coefficients for the firstplurality of taps is set to produce a first bandpass regionapproximately centered at the oscillation frequency of the analog signaland having a first bandwidth tuned for filtering a first pure tonecomponent of the set of pure tone components; and converting, by asecond digital BPF circuit of the digital BPF circuit, the seconddigital signal of the n second digital signals into a second outbounddigital signal of the plurality of outbound digital signals, wherein thesecond digital BPF circuit includes a second plurality of taps having asecond plurality of coefficients, wherein the second plurality ofcoefficients for the second plurality of taps is set to produce a secondbandpass region approximately centered at the oscillation frequency ofthe analog signal and having a second bandwidth tuned for filtering asecond pure tone component of the set of pure tone components.
 6. Themethod of claim 5 further comprises: selecting, by a processing moduleof the analog to digital conversion circuit, the first and second puretone components; and computing, by a coefficient processer of thedigital BPF circuit, the first and second plurality of coefficientsbased on the selected first and second pure tone components.
 7. Themethod of claim 1 further comprises: the first data rate frequency is2¹⁷ times the third data rate frequency; and the second data ratefrequency is 2¹² times the third data rate frequency.
 8. The method ofclaim 1 further comprises: the first data rate frequency includes afirst bit rate per interval; the second data rate frequency includes asecond bit rate per interval; and the third data rate frequency includesa third bit rate per interval.
 9. The method of claim 1 furthercomprises: determining, by a processing module of the analog to digitalconversion circuit a plurality of impedance values from the plurality ofoutbound digital signals, wherein a first outbound digital signal of theplurality of outbound digital signals is a vector having real andimaginary components.
 10. The method of claim 1, wherein the bandwidthtuned for filtering the pure tone component is less than 0.05 of channelspacing, and wherein the channel spacing is the bandwidth between twopure tone components of the set of pure tone components.
 11. A methodcomprises: converting, by n analog to digital converter (ADC) circuitsof a touch screen display, n analog signals into n first digital signalshaving a first data rate frequency, wherein an analog signal of the nanalog signals includes an oscillation frequency and a set of pure tonecomponents, wherein n is an integer greater than 1, and wherein thetouch screen display includes n electrodes; converting, by n digitaldecimation filtering circuits of the touch screen display, the n firstdigital signals into n second digital signals having a second data ratefrequency; converting, by n digital bandpass filter (BPF) circuits ofthe touch screen display, the n second digital signals into a pluralityof outbound digital signals having a third data rate frequency, whereina digital BPF circuit of the n digital BPF circuits includes a pluralityof taps having a plurality of coefficients, wherein the plurality ofcoefficients for the plurality of taps is set to produce a bandpassregion approximately centered at the oscillation frequency of the analogsignal and having a bandwidth tuned for filtering a pure tone componentof the set of pure tone components, wherein the first data ratefrequency is a first integer multiple of the third data rate frequency,wherein the second data rate frequency is a second integer multiple ofthe third data rate frequency, and wherein the first data rate frequencyis greater than the second data rate frequency; determining, by aprocessing module of the touch screen display, a plurality of impedancevalues from the plurality of outbound digital signals, wherein a firstoutbound digital signal of the plurality of outbound digital signals isa vector having real and imaginary components; and interpreting, by theprocessing module, the plurality of impedance values as touch data ofthe touch screen display.
 12. The method of claim 11, wherein the firstdata rate frequency is a third integer multiple of the second data ratefrequency.
 13. The method of claim 11, wherein a first digital signal ofthe n first digital signals is a 1-bit digital data stream at the firstdata rate frequency.
 14. The method of claim 11 further comprises:converting, by a first digital BPF circuit of the digital BPF circuit, asecond digital signal of the n second digital signals into a firstoutbound digital signal of the plurality of outbound digital signals,wherein the first digital BPF circuit includes a first plurality of tapshaving a first plurality of coefficients, wherein the first plurality ofcoefficients for the first plurality of taps is set to produce a firstbandpass region approximately centered at the oscillation frequency ofthe analog signal and having a first bandwidth tuned for filtering afirst pure tone component of the set of pure tone components; andconverting, by a second digital BPF circuit of the digital BPF circuit,the second digital signal of then second digital signals into a secondoutbound digital signal of the plurality of outbound digital signals,wherein the second digital BPF circuit includes a second plurality oftaps having a second plurality of coefficients, wherein the secondplurality of coefficients for the second plurality of taps is set toproduce a second bandpass region approximately centered at theoscillation frequency of the analog signal and having a second bandwidthtuned for filtering a second pure tone component of the set of pure tonecomponents.
 15. The method of claim 14 further comprises: selecting, bythe processing module, the first and second pure tone components; andcomputing, by a coefficient processer of the digital BPF circuit, thefirst and second plurality of coefficients based on the selected firstand second pure tone components.
 16. The method of claim 15 furthercomprises: selecting, by the processing module, the first pure tonecomponent based on a known frequency; and selecting, by the processingmodule, the second pure tone component based on a region of interest ofthe touch screen display.
 17. The method of claim 11, wherein the puretone component corresponds to one of: a self capacitance of an electrodeof then electrodes; a pen capacitance of the electrode; and a mutualcapacitance of the electrode.
 18. The method of claim 11 furthercomprises: the first data rate frequency is 2¹⁷ times the third datarate frequency; and the second data rate frequency is 2¹² times thethird data rate frequency.
 19. The method of claim 11 further comprises:the first data rate frequency includes a first bit rate per interval;the second data rate frequency includes a second bit rate per interval;and the third data rate frequency includes a third bit rate perinterval.
 20. The method of claim 11, wherein the bandwidth tuned forfiltering the pure tone is less than 0.05 of channel spacing, whereinthe channel spacing is the bandwidth between two pure tone components ofthe set of pure tone components.